Beamwidth-Reconfigurable Circularly Polarized Slot Antenna Based on Half-Mode Substrate-Integrated Waveguide

Abstract

Beamwidth-reconfigurable antennas are useful for the intersatellite link of low earth orbit formation flying and constellation, as they prevent unauthorized satellites from eavesdropping. In this article, a circularly polarized slot array antenna based on a half-mode substrate-integrated waveguide (HMSIW) for the K-band beamwidth reconfiguration is proposed using a new radio frequency (RF) switch structure and a pair of modified −45° and +45° linearly polarized HMSIW slot arrays for the dual operation of a single-pole double-throw (SPDT)/a power divider (PD) and easy integration with other components, respectively. The RF switch structure consists of a T-junction PD, λ/4 lines, and beam lead PIN diodes with current control resistors and without a DC block circuit for low DC power consumption and size reduction. The −45°/+45° linearly polarized HMSIW slot arrays providing linear and circular polarizations (LP and CP, respectively) are operated for CP. The use of a short-circuited termination instead of dissipative termination results in easier integration with other components because the 16 radiating slots consume most of the input power. The dimension of the beamwidth-reconfigurable antenna including the bottom metal layer is 157.2 × 23.3 × 0.254 mm³ (12.5λ₀ × 1.86λ₀ × 0.0202λ₀). The RF switch for the SPDT shows the insertion losses of 1.8–2.3 and 16.7–24.2 dB and an isolation of 20.9–33.4 dB for both outputs within the 10-dB bandwidth. The RF switch for the PD has an insertion loss of 3.9–4.8 dB. The one- and two-antenna operation modes of the CP antenna provide the gains of 9.44 and 6.99 dBic, the axial ratios of 2.24 and 3.47 dB, and the horizontal beamwidths of 35.8° and 78.2°, respectively.

1. Introduction

Reconfigurable antennas have been increasingly used in wireless communication systems, owing to their selectivity for operating frequencies and radiation patterns [1,2,3,4,5,6,7,8,9,10,11]. Frequency-reconfigurable antennas can operate at different frequency bands during the operation of switches. By virtue of the use of only one element for a multi-standard system, frequency-reconfigurable antennas have the advantages of reducing the complexity and improving the size form factor for the overall system [1]. On the other hand, a beamwidth-reconfigurable antenna can help improve immunity against interference, such as noise sources and intentional jamming signals [5]. This antenna is suitable for base station applications in which interference sources and a dynamic number of subscribers exist in the area of concern simultaneously, requiring the real-time control of the beamwidth and power coverage of the antenna [5]. It can also be used for satellite communications from elliptical orbits or remote sensing from airborne platforms, in which the dynamic control of the radiation beamwidth is desired to achieve a radiation beam spot independent of the antenna altitude [12]. Furthermore, beamwidth-reconfigurable antennas are beneficial for the intersatellite link (ISL) of low earth orbit (LEO) formation flying and constellation, as they prevent unauthorized satellites from eavesdropping.

Methods for antennas with beamwidth reconfigurations are classified into two categories: mechanical approaches and electronic control methods. Mechanical approaches, such as defocusing the feed system [13] or the mechanical actuation of the director/reflector [14,15], have been proposed. Structures configured by these methods are bulky and vulnerable to vibrations, misalignment, and hysteresis effects. Electric control methods have been studied using radio frequency (RF) switches, including field-effect transistors (FETs) [16], microelectromechanical systems (MEMS) [17], PIN diodes [1,2,18], and varactor diodes [5,12]. Figure 1 shows the antenna beam pattern according to an electrically controlled switch state.

FET switches have the disadvantages of nonlinear distortions and high switching losses. MEMS switches have the disadvantages of a high actuation voltage and complicated fabrication process. PIN diodes in the ON state consume considerable DC power up to tens or even hundreds of milliwatts. Varactor diodes are not suitable for the K-band or millimeter-wave band applications due to their packaging parasitic inductance. In general, the PIN diode is used for connecting or disconnecting two transmission lines for RF signal path. Even when beam lead PIN diodes with low parasitic inductance are applied to the series type diode networks, excellent on/off characteristics related to the connection and disconnection operations are not to be expected for the K-band or millimeter-wave band. A new RF switch structure is required for an ISL with a K-band beamwidth-reconfigurable antenna to prohibit unauthorized satellites from eavesdropping. Furthermore, the RF switch with a low DC power consumption of a few milliwatts is beneficial for energy saving.

This article presents a circularly polarized (CP) slot antenna for the K-band beamwidth reconfiguration introducing an RF switch with a T-junction power divider (PD), λ/4 lines, beam lead PIN diodes, current control resistors, and no DC block circuit, and −45°/+45° linearly polarized (LP) half-mode substrate-integrated waveguide (HMSIW) slot arrays with short-circuited termination and 16 radiating slots. The overall CP slot antenna was designed on a microwave substrate RO4350B with a thickness of 10 mil and a relative dielectric constant ${\mathsf{\epsilon }}_{\mathrm{r}}$ of 3.48.

2. Design

2.1. RF Switch

An RF switch for the K-band has been studied using a light-controllable photoconductive element made by intrinsic silicon wafer [19]. The RF switch based on the photoconductive element is expensive and consumes a DC power of 200 mW. An RF switch using a beam lead PIN diode and switchable HMSIW unit operating in the frequency band from 4.6 to 5.3 GHz has been presented [18]. Its structure can be applied to K-band applications; however, it requires sufficient space for the DC circuits and consumes a DC power of 90 mW.

In this article, an RF switch using beam lead PIN diode networks of excellent on/off characteristics and a DC power consumption of a few milliwatts was proposed at the K-band. A schematic of the proposed RF switch operating as a single-pole double-throw (SPDT) and PD for beamwidth reconfiguration is shown in Figure 2. The proposed SPDT/PD for the K-band consists of a T-junction PD, λ/4 lines, and diode networks. In this design, the beam lead PIN diodes DSM8100-000 manufactured by the Skyworks Company (Irvine, CA, USA), which can operate up to 25 GHz, were used [20,21]. The diode could be modeled as a resistance of 4 Ω for the ON state and as a parallel circuit with a capacitance of 0.03 pF and a resistance of 20 kΩ for the OFF state [21]. Each diode network consisted of a K-band short-circuited radial stub, PIN diode, and transmission line, and was controlled by a DC bias network containing a current control resistor of 2 kΩ for low DC power consumption. The designed shut type diode network with the effect of the parasitic inductance reduced by the transmission line was superior on/off characteristics to the series type diode network for the K-band. The cathode of the PIN diode was placed toward point A (A′) in Figure 2 and connected to a ground via, which resulted in a reduction in the insertion loss and circuit size, owing to the absence of a DC block circuit. The OFF and ON states of the diode were controlled with the voltages of 0 V and 3 V, respectively. The diode network at point A (A′) in Figure 2 served as an open or a short circuit according to the control voltage of 0 V or 3 V, respectively, and each impedance at point A toward antenna 1 (2) was transformed by a λ/4 line between A-B (A′-B), denoted as TL2 in Figure 2, into 50 Ω or $\infty$ at point B, respectively. Therefore, the operational state of the antenna could be determined by the state of the diode network connected to the antenna feed line. The proposed RF switch for antenna feed control served as the SPDT with the operation state of “Ant1 ON, Ant2 OFF” or “Ant1 OFF, Ant2 ON” and the PD with the operation state of “Ant1 ON, Ant2 ON”. A λ/4 impedance transformer with a characteristic impedance of 41.5 Ω, denoted as TL1 in Figure 2, was adopted, satisfying the 10-dB impedance for both SPDT and PD operations.

2.2. HMSIW CP Slot Antenna

Currently, substrate-integrated waveguide (SIW) technology is promising for millimeter-wave applications by virtue of its low loss and ease of fabrication with RF circuits. CP slot antennas based on the SIW technology have been presented [22,23,24]. HMSIW structures have been used for size reduction, which was achieved by bisecting a conventional SIW along a fictitious magnetic wall. A slot antenna based on the HMSIW technology and leaky waves has been presented for linear and circular polarizations (LP and CP, respectively) [25]. In HMSIW, the quasi-TE_(m-0._5),0 modes could propagate, and the dominant quasi-TE₀._5,0 mode was addressed (m = 1) for slot antennas [26]. A 45°-inclined HMSIW slot analysis model is illustrated in Figure 3. Both series impedance Z₁ and parallel impedance Z₂ existed because the slot was off the centerline (bisection line for HMSIW) of the SIW [27]. From the equivalent circuit model, the calculation formulas for Z₁ and Z₂ are expressed as follows [27]:

$$\frac{Z_1}{Z_0}=\frac{1+S_{11}-S_{21}}{1-S_{11}},$$

(1)

$$\frac{Z_2}{Z_0}=\frac{S_{21}}{1-S_{11}-S_{21}}.$$

(2)

The radiation power rate can be calculated as follows:

$$\frac{P_r}{P_{in}}=1-{\left|S_{11}\right|}^2-{\left|S_{21}\right|}^2,$$

(3)

where $P_r$ and $P_{in}$ are the radiation and input powers, respectively.

In this study, an HMSIW slot antenna providing −45° and +45° LPs, left-handed CP (LHCP), and right-handed CP (RHCP) with input ports on the same side for easier integration with other components was introduced. The proposed HMSIW slot antenna is shown in Figure 4. The antenna was constructed using −45° and +45° LP antennas, each of which had 16 radiating slots and a short-circuited termination. If the slot spacing of a slot array is not just half of the guided wavelength and a matched termination is placed at the end, a leaky-wave array may be obtained [25]. The proposed slot array antenna, whose slot spacing was not just half of the guided wavelength, operated as a leaky-wave array, even though it used a short-circuited termination because the array with radiating slots was long enough to allow the power at the termination to become a small fraction of the radiated power. The solution using short-circuited termination with the residual power at the termination being less than 2% of the radiated power was virtually equivalent to the solution using matched termination [22]. The HMSIWs of both antennas faced each other by sharing inner vias for the metallic walls of the HMSIW. An additional row of shunt vias (edge via) near the open edge was responsible for the improvement in the return loss [25] and the reduction in antenna coupling between the two adjacent arrays. Each slot of the −45° LP antenna was placed at the middle of each adjacent slot of the +45° LP antenna along the z-axis for a negligible effect on the radiation patterns. For speculation of the CP, the HMSIW slot antenna was realized using a Branch-line coupler for a circular polarizer and a line with a phase angle of 180° for port 2 (Figure 4). The CP mechanism was confirmed by vector current distribution in simulation [28,29]. The surface current distributions of the antenna for the LHCP and RHCP were analyzed from Figure 5 and Figure 6, respectively, with the variation of the phase angle at 0°, 90°, 180°, and 270° at the frequency of 23.9 GHz. The two ports of the antenna were excited with the phase differences of −90° and +90° for both CP. For the LHCP in Figure 5, the maximum current vectors were $+$x$+$z, $-$x$+$z, $-$x$-$z, and $+$x$-$z directions (Figure 4) at the phase angles of 0°, 90°, 180°, and 270°, respectively. For the RHCP in Figure 6, the maximum current vectors were $+$x$-$z, $-$x$-$z, $-$x$+$z, and $+$x$+$z directions at the phase angles of 0°, 90°, 180°, and 270°, respectively.

3. Results

3.1. RF Switch

The fabricated RF switch for the beamwidth reconfiguration is shown in Figure 7. The S-parameters of the designed RF switch according to the switch state of the antenna are shown in Figure 8. The switch states of ports 2 and 3 for the RF switch correspond to the states of antennas 1 and 2, respectively (Figure 2). The SPDT mode for antenna 1 and the PD for antennas 1 and 2 are shown. The designed RF switch for the SPDT/PD provides a 10-dB impedance bandwidth (BW) of 23.34–24.60 GHz and a fractional bandwidth (FBW) of 5.3% at a center frequency of 23.9 GHz. The RF switch serves as a SPDT with the insertion losses of 1.8–2.3 and 16.7–24.2 dB and an isolation of 20.9–33.4 dB for both outputs within the 10-dB BW. The RF switch acts as a PD with an insertion loss of 3.9–4.8 dB within the 10-dB BW. The DC power consumption was 3 mW (3 V and 1 mA) when each diode was switched on. The performance of the proposed RF was compared with that of published RF switches in

Table 1. Performance comparison of the proposed RF switch with published RF switches.

		[30]	[31]	[18]	[19]	This Work
Function	Center Freq. (GHz)	2.14	2.5	4.95	24	23.9
	FBW (%)	~14	8		7.5	5.3
Power divider	R.L.@port1 (dB)	24.3	-	14.5	-	10.6–27.5
	R.L.@port2 (dB)	25	-	7.4	-	10.0–20.3
	I.L. (dB)	3.3	-	4.35	-	3.9–4.8
SPDT @Port2 (ON state)	R.L.@port1 (dB)	10	10	15	>10	10–13.9
	R.L.@port2 (dB)		-	13	-	10–38.6
	I.L.@port1-2 (dB)	1.8	2	1.35	>1.9	1.8–2.3
	I.L.@port1-3 (dB)	10.7	12	23	-	16.7–24.2
	Isolation (dB)		-	27	25	20.9–33.4
DC power consumption (mW)		-	-	90	200	3
Note.		Microstrip line /PIN diode	SIW /PIN diode	SIW /PIN diode	SIW /photo- conductive element	Microstrip line /PIN diode

. The designed K-band RF switch demonstrated a good performance compared with the other switches. Moreover, the DC power consumption was drastically reduced.

3.2. HMSIW CP Slot Antenna

The fabricated HMSIW RHCP slot array antenna is shown in Figure 9. The dimension including the bottom metal layer was 143.8 $\times$ 10.1 $\times$ 0.254 mm³ ($11.46{\lambda }_0\times 0.80{\lambda }_0\times 0.0202{\lambda }_0)$. It comprised a pair of −45° and +45° LP antennas, a circular polarizer, and a 180° transmission line. As depicted in Figure 10, the circular polarizer was realized using two parallel-connected 0201 flip chip resistors of 100 Ω for the 50 Ω termination at port 4, to reduce the parasitic effect on the K-band application. The input signal was excited at port 1 and divided by the designed circular polarizer into two signals with the relative phases of 0° and 90° at ports 2 and 3, respectively (Figure 9 and Figure 10). Both signals at ports 2 and 3 propagated on the transmission lines with the relative phases of 0° and 180°, respectively (Figure 9), and excited the −45° and +45° LP antennas with the relative phases of 0° and 270°, respectively. Consequently, an RHCP radiation pattern was formed. When port 1 was 50 Ω terminated, and port 4 was used as the input port, an LHCP radiation pattern was generated.

The fabricated circular polarizer provided the gain imbalance and isolation of less than 0.4 dB and more than 21.4 dB, respectively, and the HMSIW RHCP slot array antenna has a wide 15-dB BW, as shown in Figure 11. The radiation patterns of the HMSIW RHCP slot array antenna were measured with Friis radio link formula and compact antenna test range and shown in Figure 12. Friis radio link formula is expressed as follows:

$$P_r =\frac{G_t{G}_r{\lambda }^2}{{\left(4\pi R\right)}^2}{P}_t,$$

(4)

where $P_r$, $P_t$, $G_t$, $G_r$, and $\lambda$ are the received power, transmit power, transmit antenna gain, receive antenna gain, operating wavelength, and a distance between both antennas, respectively. The simulated results are based on an ideal circular polarizer. The HMSIW RHCP slot array antenna provides a gain of 8.57 dBic, an axial ratio (AR) of 3.4 dB, a half-power beamwidth (HPBW) of 5°, a side-lobe level of 8.4 dB, and a co-to-cross-pol ratio of 12.3–20.1 dB within the HPBW at the frequency of 23.9 GHz in measurement. The discrepancies between the simulated and measured gains would be mainly attributed to the performance of the fabricated circular polarizer, manufacturing tolerances, and conductor and dielectric losses in the K-band.

3.3. Conjunction for Beamwidth-Reconfigurable Antenna

The K-band RF switch was applied to the HMSIW RHCP slot array antenna to realize beamwidth reconfiguration. Figure 13 illustrates the fabricated beamwidth-reconfigurable HMSIW RHCP slot array antenna. The dimension including the bottom metal layer was 157.2 $\times$ 23.26 $\times$ 0.254 mm³ ($12.5 {\lambda }_0\times 1.86{\lambda }_0\times 0.0202{\lambda }_0)$. The operating modes of the antenna are listed in

Table 2. Operation modes of the beamwidth-reconfigurable antenna.

	Left Antenna	Right Antenna
Mode 1	ON	ON
Mode 2	ON	OFF
Mode 3	OFF	ON
Mode 4	OFF	OFF

. The horizontal and vertical radiation patterns of the four modes are shown in Figure 14. The simulated results were based on an ideal switch and circular polarizer. The gain, AR, efficiency, and reflection coefficient of the designed antenna for each mode against the frequency are shown in Figure 15. The measured gains of modes 1, 2, 3, and 4 were 9.44, 6.09, 8.46, and 0.57 dBic, respectively, at 23.9 GHz. The measured ARs of modes 1, 2, and 3 were 2.24, 3.47, and 3.50 dB, respectively, at 23.9 GHz. As expected, the horizontal beamwidth for mode 1 became narrower than that for modes 2 or 3. The measured radiation efficiencies of modes 1, 2, and 3 were 18.93%, 18.01%, and 18.79%, respectively, at 23.9 GHz. The discrepancies between the simulated and measured results would be mainly attributed to the performance of the fabricated RF switch and circular polarizer, manufacturing tolerances, and conductor and dielectric losses in the K-band. The gain and efficiency of the antenna can be improved when a substrate with low loss tangent, such as an RT/duroid 5880, is used.

4. Conclusions

This article proposed a beamwidth-reconfigurable HMSIW CP slot antenna for the K-band with an RF switch and HMSIW CP slot antenna. The K-band RF switch, operating as an SPDT/PD and consisting of a T-junction PD, λ/4 lines, and beam lead PIN diodes, demonstrated a good performance in terms of insertion loss, isolation, and DC power consumption, owing to the excellent on/off characteristics of the beam lead PIN networks and the current controls of the DC bias networks. The HMSIW CP slot array antenna included a circular polarizer and a pair of −45° and +45° LP HMSIW slot arrays. The LP HMSIW slot array had the advantage of easier integration with other components, owing to the inclusion of 16 slots and short-circuited termination. The −45°/+45° LP HMSIW slot arrays provided LP and CP, respectively, and the CP antenna was applied to the RF switch. The beamwidth reconfiguration of the antenna was confirmed through measurements. It is expected that the proposed antenna structure will be used for the ISL of the LEO formation flying and constellation, in order to prevent unauthorized satellites from eavesdropping.