Design of High-Performance Driving Power Supply for Semiconductor Laser

Abstract

High power semiconductor laser is a kind of photoelectric device with high efficiency and high stability, the performance of its drive system directly affects its output characteristics and service life. In order to solve the problems of stability and robustness of the output power of the semiconductor laser, a semiconductor laser driving power supply with high efficiency, low ripple and strong anti-interference ability was developed. In this paper, the topology of the LCC resonant converter is adopted (LCC refers to the type of resonant converter, because its resonator is composed of an inductor L and two capacitors C, it is called LCC resonant converter). The power supply adopts full-bridge LCC resonant power topology. Firstly, a mathematical model is established to analyze the relationship between LCC resonator parameters and output current gain. Secondly, an LCC resonator parameter design method is proposed to reduce the current stress of components, and the variable frequency phase shift (PFM-PWM) composite control strategy and linear active disturbance rejection control (LADRC) algorithm are proposed, which not only ensures the zero voltage (ZVS) conduction of MOS (Metal-Oxide-Semiconductor) tube, but also reduces the on-off loss of MOS tube. The PFM-PWM composite control strategy and LADRC algorithm not only improve the power efficiency of the drive power supply, suppress the output current ripple, but also ensure that the output current of the drive power supply is stable when the input voltage, load and parasitic parameters of the circuit change. Finally, the simulation and experimental results show that the power supply can be continuously adjustable in the output current range of 0–40 A, the current ripple is less than 0.8%, and the working efficiency is up to 92%. It has the characteristics of high stability, small ripple, high efficiency, low cost and good robustness.

1. Introduction

High power semiconductor lasers have many advantages such as light weight, high efficiency, small size, high reliability and long service life, and are widely used in production, medical, aerospace, national defense and scientific research and other fields [1,2]. The extensive utilization of the high-power semiconductor laser is closely linked to the ongoing enhancement and advancement of its driving power supply [3,4]. The laser diode in the core device of the semiconductor laser pump source exhibits low resistance to electrical shock, making it susceptible to significant deviations in output optical power due to even slight current fluctuations. Additionally, transient current spikes in the drive power circuit can lead to damage of the laser diode and negatively impact its operational lifespan [5,6]. The laser diode in the core device of the semiconductor laser pump source exhibits limited resistance to electrical shock, and the performance of the laser power supply directly determines the overall performance of the entire laser system as the output power of the semiconductor laser continues to increase. Therefore, it is essential for the semiconductor laser drive power supply to possess high stability, high power density, low ripple, robustness, and adaptability to complex environmental applications [7].

Based on the load characteristics of semiconductor lasers, constant current mode is the preferred operating mode for their driving power supply. Both domestic and foreign scholars have conducted extensive research on designing driving powers for semiconductor lasers to meet various application requirements [8,9,10].

International research on semiconductor lasers commenced earlier, exemplified by the QCL (the code for a series of power supplies) driver power supply from Wavelength company and the LDC202C power supply developed by Thorlabs company [11,12]. These devices offer a precision of up to 0.05 mA, along with flexible adjustability and high stability; however, their cost is relatively high, making widespread adoption challenging. The researchers at Kassel University in Germany, Zhang et al. [13], employed an oscillator circuit to generate high-frequency pulses for precise control of semiconductor lasers, thereby introducing a novel approach to drive semiconductor lasers. Lu Yi et al., from the University of Electronic Science and Technology of China [14], proposed a double closed-loop constant current drive strategy. They utilized the LM25117 chip, which is based on a synchronous rectifier Buck power supply, to effectively regulate the steady-state loss of MOSFET in the constant current circuit. As a result, the steady-state operating efficiency of the power supply reached 88%, thereby significantly enhancing its overall power efficiency. The utilization of the LM2517 control chip exhibits limited flexibility and inadequate anti-interference capability. However, the LM2517 control chip lacks flexibility and has poor anti-interference ability. The researchers at Huazhong University of Science and Technology, led by Wang Chao et al. [15], developed a single-stage conversion switching DC drive power supply with a capacity of up to 2 kW using the parallel interleaving technology of single-tube positive cataclastic switching. This design achieved an impressive power efficiency of 85% and enabled dynamic regulation of output current from 0–100 A. However, precise control of the current balance for each individual tube converter is essential due to the large volume of the positive shock converter and the implementation of parallel interleaving technology, resulting in a complex control circuit and significant limitations in its application. Zhong Xulang et al. [16] from Dazu Laser Technology proposed a power supply structure that combines MOS tubes and operational amplifiers, enabling real-time acquisition of power data, multi-channel current output, and laser power error control within ±2%. But the use of operational amplifier MOS tube will cause the system power loss to be too large, the control flexibility to be poor, and the overall efficiency of the power supply to be low. The researchers at Yanshan University, led by Zhao Qinglin et al. [17,18], have developed a structure that combines the LCC resonant converter capacitor charging circuit with a pulse current circuit. This innovative design enables the pulse current output to reach an amplitude of 80 A, allowing for wide voltage output range and ensuring high flexibility and stability.

In order to address the limitations of the aforementioned semiconductor laser drive power supply, this paper adopts a topology consisting of full-bridge LCC resonant transformation, variable frequency phase shift (PFM-PWM) composite control strategy, and a linear active disturbance rejection control (LADRC) algorithm. This approach not only enhances the dynamic response speed, stability, and efficiency of the semiconductor laser drive power supply but also enables a wide range of input/output capabilities.

2. The Composition of a Power System

The proposed structure of the semiconductor laser power supply system is illustrated in Figure 1, comprising primarily of an LCC resonant converter module, a controller module, a current acquisition module, and an auxiliary power supply module.

The LCC resonant converter module, depicted in Figure 1, serves as the cornerstone of the power supply system by delivering a constant current. It encompasses a full-bridge inverter network, an LCC resonant network, a full-bridge rectifier network, and an LC filter network. The current acquisition module primarily samples the output current into the controller module, establishing a closed-loop feedback control mechanism. The controller module utilizes the appropriate algorithm to calculate the sampling current, thereby generating the signal that drives the MOS tube. It also assists in supplying power to the controller, drive circuit, and current sampling circuit.

2.1. Mathematical Model Analysis of LCC Resonant Converter

The topology of the LCC resonant converter is illustrated in Figure 2. From a topological perspective, the LCC resonant converter exhibits not only the characteristics of a series resonant converter for DC component isolation and transformer protection but also possesses wide-range input/output regulation capabilities akin to those of a parallel resonant converter. Consequently, it demonstrates excellent constant current source characteristics and robust output short-circuit resistance [19,20]. In order to optimize the volume of resonant converter, reduce the circuit current and circuit equivalent capacitance, increase the characteristic impedance of resonant network, and reduce the current stress borne by the converter, the parallel resonant capacitor ($C_P$) is designed on the secondary side of the transformer, so that the parallel resonant capacitor ($C_P$) and series resonant inductor ($L_r$) become parasitic parameters that cannot be ignored by the transformer under high frequency operation. It has a good internal short-circuit protection function.

Because the mathematical model of the LCC resonant converter is nonlinear, it is not conducive to the analysis of its working state, so it is necessary to linearize its nonlinear terms. When the frequency of the small-amplitude disturbed signal is much lower than the switching frequency of the system, the whole resonant converter can be considered as a quasi-steady-state system. In order to simplify the analysis, extended description function analysis [21,22] and linear differential equation theory are used to approximate the nonlinear terms, and the mathematical model of the steady-state system is obtained:

$$L_r(\frac{d{i}_{rs}}{dt}-{\omega }_s{i}_{rc})+v_{c_{s}s}+n{v}_{C_p^{\prime }s}=\frac{4v_g}{\pi }\sin (\frac{d}{2})$$

(1)

$$L_r(\frac{d{i}_{rc}}{dt}+{\omega }_s{i}_{rs})+v_{{\mathrm{c}}_{s}c}+n{v}_{C_p^{\prime }c}=0$$

(2)

$$i_{rs}=C_s(\frac{d{v}_{c_{s}s}}{dt}-{\omega }_s{v}_{{\mathrm{c}}_{s}c})$$

(3)

$$i_{rc}=C_s(\frac{d{v}_{{\mathrm{c}}_{s}c}}{dt}+{\omega }_s{v}_{c_{s}s})$$

(4)

$$C_{\mathrm{p}}^{\prime }(\frac{\mathrm{d}{v}_{C_{\mathrm{p}}^{\prime }s}}{\mathrm{d}t}-{\mathrm{\omega }}_{\mathrm{s}}{v}_{C_{\mathrm{p}}^{\prime }c})+\frac{4i_{L_{\mathrm{f}}}}{\mathrm{\pi }{A}_{\mathrm{p}}}{v}_{C_{\mathrm{p}}^{\prime }s}=n{i}_{\mathrm{rs}}$$

(5)

$$C_{\mathrm{p}}^{\prime }(\frac{\mathrm{d}{v}_{C_{\mathrm{p}}^{\prime }c}}{\mathrm{d}t}+{\mathrm{\omega }}_{\mathrm{s}}{v}_{C_{\mathrm{p}}^{\prime }s})+\frac{4i_{L_{\mathrm{f}}}}{\mathrm{\pi }{A}_{\mathrm{p}}}{v}_{C_{\mathrm{p}}^{\prime }c}=n{i}_{\mathrm{rc}}$$

(6)

$$L_{\mathrm{f}}\frac{\mathrm{d}{i}_{L_{\mathrm{f}}}}{\mathrm{d}t}+i_{L_{\mathrm{f}}}{r}_{\mathrm{c}}^{\prime }+(1-\frac{r_{\mathrm{c}}^{\prime }}{R_{\mathrm{L}}})v_{c_{\mathrm{f}}}=\frac{2}{\mathrm{\pi }}\sqrt{v_{C_{\mathrm{p}}^{\prime }s}^2+v_{C_{\mathrm{p}}^{\prime }c}^2}$$

(7)

$$\frac{R_{\mathrm{L}}}{r_{\mathrm{c}}+R_{\mathrm{L}}}{C}_{\mathrm{f}}\frac{\mathrm{d}{v}_{C_{\mathrm{f}}}}{\mathrm{d}t}+\frac{1}{R_{\mathrm{L}}}{v}_{C_{\mathrm{f}}}=i_{L_{\mathrm{f}}}+i_{\mathrm{o}}$$

(8)

The output voltage $v_o$ is as follows:

$$v_0=R_{\mathrm{L}}{i}_{\mathrm{o}}=i_{L_{\mathrm{f}}}{r}_{\mathrm{c}}^{\prime }+(1-\frac{r_{\mathrm{c}}^{\prime }}{R_{\mathrm{L}}})v_{c_{\mathrm{f}}}$$

(9)

i_o is the output current; v_o the output voltage; i_rs and i_rc are amplitude of sinusoidal component and cosine component of resonant current i_r, respectively; $v_{c_{\mathrm{s}}\mathrm{s}},v_{{\mathrm{c}}_{\mathrm{s}}\mathrm{c}}$ are the amplitude of the sinusoidal and cosine components of the series resonant capacitor voltage $v_{c_s}$, respectively; $v_{C_{\mathrm{p}}^{\prime }\mathrm{s}},v_{C_{\mathrm{p}}^{\prime }\mathrm{c}}$ are the amplitude of the sinusoidal and cosine components of the series resonant capacitor voltage $v_{C_{\mathrm{p}}^{\prime }}$, respectively; ω_s is the operating angular frequency; n indicates the ratio of turns of the transformer; v_g indicates the input voltage; d represents duty cycle; i_Lf represents the current through the filtered inductor L_f; v_Cf represents the voltage of filter capacitor C_f. r_c represents the ESR of the filter capacitance; R_L indicates the resistance of the load.

Solve Equations (1) through (9), The voltage gain of the open loop DC point of the LCC resonant converter is obtained:

$$G_V=\frac{V_{\mathrm{o}}}{V_{\mathrm{g}}}=\frac{{\omega }_{\mathrm{s}}{R}_{\mathrm{L}}{C}_{\mathrm{s}}\sin (\frac{d}{2})}{n\sqrt{{(1-{\omega }_{\mathrm{s}}^2{L}_{\mathrm{r}}{C}_{\mathrm{e}})}^2+{({\omega }_{\mathrm{s}}{R}_{\mathrm{e}}(C_{\mathrm{p}}+C_s)(1-{\omega }_{\mathrm{s}}^2{L}_{\mathrm{r}}{C}_{\mathrm{e}}))}^2}}$$

(10)

$C_{\mathrm{e}}=\raisebox{1ex}{$C_{\mathrm{s}}{C}_{\mathrm{p}}$}\!\left/ \!\raisebox{-1ex}{$(C_{\mathrm{p}}+C_{\mathrm{s}})$}\right.$ represents the equivalent capacitance of the resonant network.

According to the relationship between voltage gain and current gain, the current gain can be obtained by normalizing Equation (10):

$$G_I=\frac{I_{\mathrm{o}}}{I_{\mathrm{i}}}=G_V{Q}_{\mathrm{L}}=\frac{Q_{\mathrm{L}}\sin (\frac{d}{2})}{n\sqrt{{(1+A)}^2{(1-\omega )}^2+Q_{\mathrm{L}}^2{(\omega -{\omega }^{-1}\frac{A}{1+A})}^2}}$$

(11)

ω = ω_s/ω_r, ω_r is the angular frequency of the resonant network, ${\omega }_{\mathrm{r}}=\sqrt{\raisebox{1ex}{$1$}\!\left/ \!\raisebox{-1ex}{$L_{\mathrm{r}}{C}_{\mathrm{e}}$}\right.}$, $Q_L=\raisebox{1ex}{$Z_{\mathrm{r}}$}\!\left/ \!\raisebox{-1ex}{$R_{\mathrm{e}}$}\right.$, $Z_{\mathrm{r}}=\sqrt{\raisebox{1ex}{$L_{\mathrm{r}}$}\!\left/ \!\raisebox{-1ex}{$C_{\mathrm{e}}$}\right.}$, Z_r indicates the characteristic impedance.

It can be seen from Equation (11) that when ω = 1, the LCC resonant converter exhibits constant current characteristics, and the output current is independent of the load, but only related to the input voltage and resonance parameters. When the LCC resonant converter works in this mode, the output current can remain unchanged even if the semiconductor laser load changes. In order to study the influence of current gain G_I on Q_L, A, d and n, assuming A = n = 1 and d = π, the change curve of current gain with different values of Q_L and A is drawn, as shown in Figure 3.

As can be seen from Figure 3a, when the QF value increases continuously, the value of the current gain also increases. When the load is open (limQ_L→∞), the voltage gain G_I will increase rapidly and tend to infinity, which will damage the power system in serious cases. When ω = 1, the current gain is independent of the load, showing constant current characteristics; As can be seen from Figure 3b, when the capacitance ratio (A value) increases, the current gain will change in the same direction as the A value, that is, with the increase in the A value, and when ω > 1, the smaller A, the more gradual the gain change which was mentioned above.

The values of duty cycle (d) and coil turns (n) of the transformer also affect the LCC resonant converter, and the gain influence curve of its output current and voltage is shown in Figure 4. The above two graphs are the relationship curves of duty cycle (d) and frequency ratio (ω) with voltage gain and current gain. The X-axis is the duty cycle, and the Y-axis is the voltage gain and the current gain, respectively. The following two graphs are the relationship curves of the transformer turn ratio (n) and frequency ratio (ω) with voltage gain and current gain. The X-axis is the ratio of turns, and the Y-axis is the voltage gain and the current gain, respectively.

According to the influence curves of d and n values on current and voltage gain, it can be seen that the voltage/current gain is affected by the duty cycle d, the number of turns of the transformer n, and the frequency ratio ω. When the duty cycle and the number of turns of the transformer are determined, the output current and voltage gain can be adjusted by adjusting the frequency ratio ω. At the same time, under the same duty cycle and transformer turns, when ω > 1, the voltage gain/current gain change is small. When ω < 1, the voltage/current gain change difference is large.

2.2. Design of LCC Resonator Parameters

Ignoring the loss in the power transmission of the LCC resonant converter, the resonant current peak I_r is obtained from the power conservation, which is expressed as follows:

$$I_{\mathrm{r}}=\frac{2\mathrm{\pi }{V}_{\mathrm{i}}{G}_I^2}{Q_{\mathrm{L}}^2{R}_{\mathrm{L}}\cos \mathrm{\phi }}$$

(12)

where, φ represents the impedance Angle.

The characteristic curve of resonant current peak I_r affected by capacitance ratio A, quality factor Q_L, impedance Angle φ and input voltage V_i is shown in Figure 5. It can be seen from the characteristic curve that when the converter current gain, load and input voltage are constant, the smaller the φ is, the smaller the value of I_r is, and the smaller the current stress of the switching tube is. Only by ensuring φ > 0 can the ZVS conduction of the MOS tube be ensured, and a sufficient margin is often left in the actual selection, generally within the range of 15° < φ < 30°. At the same time, too small a quality factor and capacitance ratio will cause the resonant current peak to be too large, and the increase in input voltage will also cause the resonant current to increase.

In order to obtain more accurate parameters of the resonant network, reduce the current stress of the resonant components, and ensure that the resonant converter can achieve ZVS conduction, the following parameter design steps are given:

(1) The peak range of resonant current is determined according to input voltage, output power and output current;

(2) Select the appropriate resonant frequency and operating frequency range;

(3) The quality factor and capacitance ratio are determined according to the resonant current peak range;

(4) The transformer ratio n is calculated according to Formula (13):

$$n=\frac{G_v{\sin }^2(\frac{\theta }{2})}{2\cos \phi }$$

(13)

In Formula (13), θ represents the conduction Angle of the rectifier tube;

(5) Finally, the LCC resonator parameters are calculated by Formulas (14)–(16).

Series resonant capacitance:

$$C_{\mathrm{s}}=\frac{\sqrt{1+A}}{2\mathrm{\pi }{n}^2{f}_{\mathrm{r}}{Q}_{\mathrm{L}}{\mathrm{R}}_{\mathrm{e}}}$$

(14)

Parallel resonant capacitance:

$$C_{\mathrm{p}}^{\prime }=n^2{C}_{\mathrm{s}}/A$$

(15)

Resonant inductance:

$$L_{\mathrm{r}}=\frac{1}{{(2\mathrm{\pi }{f}_{\mathrm{r}})}^2}\frac{A+1}{C_{\mathrm{p}}}$$

(16)

3. Power System Control Strategy and Algorithm Analysis

In order to overcome the disadvantages of the traditional control strategy, the variable frequency phase-shift composite control strategy is adopted, and its structure is shown in Figure 6.

In Figure 6, i_o* represents the given value of output current, u_d and u_f represent the control quantity of the phase and frequency of the controller output, respectively. f_s represents the carrier signal frequency, and g₁~g₄ represents the MOSFET drive signal, respectively. This control strategy enables the controller to keep the output current stable by adjusting the switching frequency and the on-angle at the same time when the input/output changes, and still realize the soft switching under the condition of narrow switching frequency and wide input voltage/wide output range, thus improving the working efficiency and the service life of the device.

When the output current is increased, the increase in switching frequency f_s will lead to the decrease in the on-angle φ. Therefore, under the combined action of f_s and φ, the converter can achieve a wide input voltage in a narrow switching frequency range and achieve steady current output and soft switching under a wide output power condition.

The traditional PID controller relies too much on the system model when designing parameters. In view of the complexity of the semiconductor laser load characteristics, the PID controller is easy to produce integral saturation and insufficient anti-interference ability, which makes the PID controller difficult to meet the design requirements. Therefore, the LADRC control algorithm is adopted, and its current loop structure is shown in Figure 7, where G_s(s) represents the control object transfer function, LESO represents the linear expansion observer, k_p represents the scale coefficient of the controller, k_d represents the differential coefficient, and b₀ represents the compensation factor. The design of this algorithm does not depend on the exact model and can observe and track the disturbance of the system in real time. It can overcome the shortcomings of the PID controller and restrain the current instability caused by multi-factor disturbance.

The total disturbance p of the system can be estimated by the linear extended observer (LESO):

$$\begin{array}{l}\dot{z}=Az+Bu+L(y-\hat{y})\\ \hat{y}=Cz\end{array}$$

(17)

$$\begin{array}{l}A=\left[\begin{array}{ccc}0& 1& 0\\ 0& 0& 1\\ 0& 0& 0\end{array}\right]\\ B={\left[\begin{array}{ccc}{b}_0& 0& 0\end{array}\right]}^{\mathrm{T}}\\ E={\left[\begin{array}{ccc}0& 0& 0\end{array}\right]}^{\mathrm{T}}\\ C=\left[\begin{array}{ccc}1& 0& 0\end{array}\right]\\ L={\left[\begin{array}{ccc}{\beta }_1& {\beta }_2& {\beta }_3\end{array}\right]}^{\mathrm{T}}\end{array}$$

(18)

where, u and y represent the input and output of the system, respectively. L is LESO gain matrix; z is LESO state variable; $\hat{y}$ is the output value estimated by the system.

The transfer function of the LCC resonant converter can be approximated as a second-order system, so it is necessary to design a third-order extended observer to observe the system state and the total disturbance.

$$\begin{array}{l}{\dot{z}}_1=z_2+l_1(y-z_1)\\ {\dot{z}}_2=z_3+l_2(y-z_1)+b_{0}u\\ {\dot{z}}_3=l_3(y-z_1)\end{array}$$

(19)

According to the bandwidth method [23], observer bandwidth ω_o, all poles of the extended observer are assigned in the left half plane, and the parameter tuning problem is simplified to the selection of observer bandwidth, and the corresponding observer gain is as follows:

$$\begin{array}{l}{l}_1=3{\omega }_{\mathrm{o}}\\ l_2=3{\omega }_{\mathrm{o}}^2\\ l_3={\omega }_{\mathrm{o}}^3\end{array}$$

(20)

The reconstructed system after disturbance compensation can be approximated as an integral series system. When the disturbance is observable, the second-order system can be reconstructed as a second-order integral series system by disturbance compensation. After disturbance compensation, the series integral system can obtain good control performance through simple linear error feedback rate control. The controller design is as follows:

$$u_{\mathrm{o}}=k_{\mathrm{p}}(r-z_1)-k_{\mathrm{d}}(\dot{r}-z_2)$$

(21)

where, r represents the given input; $\dot{r}$ represents the first derivative of a given input; u₀ indicates the controller output.

4. Simulation and Test Results

In order to verify the correctness of the semiconductor laser power supply scheme, Matlab/Simulink is used to build a semiconductor laser drive power supply simulation model. The prototype test platform with the LCC resonant converter and STM32 controller as the core is shown in Figure 8. By testing the constant current output characteristic, anti-interference characteristic, working efficiency and ripple characteristic of the semiconductor laser driving power supply, it is proved that the design is reasonable and meets the design requirements.

The output voltage of the test prototype is 10~15 V, the output current is 0~40 A adjustable, the operating frequency is 51.5 kHz, the resonant capacitor C_s is 354 nF, the resonant capacitor C′_p is 9.03 μF, the resonant inductor L_r is 42.9 μH, and the transformer turns ratio n is 3.5.

4.1. Constant Current Characteristic Test of Semiconductor Laser Power Supply

The simulation and test results of the constant current (40 A) characteristics of the semiconductor laser power supply are shown in Figure 9, where V_gs and V_ds are, respectively, the drive voltage and the drain-source voltage of the MOS tube, V_ab is the input voltage of the resonant network, I_r is the resonant current, and V_cp is the shunt capacitor voltage. It can be seen from the simulation and test results that the variable frequency phase-shift composite control method extends the input voltage/output power range of the LCC resonant converter by synchronously adjusting the switching frequency and duty ratio, narifies the switching frequency range, and effectively reduces the current and voltage stress of the switching tube. The output current ripple coefficient is <0.8%, and the output voltage ripple coefficient is <0.45%. It can realize ZVS conduction under any power input voltage, effectively reduce the loss of power tube, and improve the performance of LCC resonant converter.

4.2. Experiment on Anti-Interference Characteristics of Semiconductor Laser Power Supply

The simulation and test results of anti-interference characteristics of semiconductor laser power supply are shown in Figure 10. When t = 0.02~0.03 s, the input voltage V_in = 60 V changes to V_in = 80 V, and when t = 0.04~ 0.05 s, the load switches from full load to light load. According to the simulation and test results, the output current of the PID algorithm has a fast response speed, a large overjump, a large error and a large current ripple. When the input voltage or load is transformed, the ripple of the current and voltage increase, and the current and voltage remain at the set value. Therefore, the LADRC algorithm has a good performance in anti-interference.

4.3. Power Efficiency Characteristic Test of Semiconductor Laser

The simulation and test outcomes of the semiconductor laser power supply efficiency, as illustrated in Figure 11, reveal that the conventional frequency conversion control exhibits limited output adjustment capability and stability, leading to an overall low power supply efficiency. Consequently, only traditional phase shift control and composite control with frequency conversion phase shift are subjected to testing and analysis. Based on the simulation and test results, various control strategies yield differing power efficiencies; specifically, the phase-shifting control strategy induces hard switching states in the converter switch, thereby increasing switching losses and reactive power losses of the resonant converter. By implementing the inverter phase-shifting compound control under full load conditions, a maximum power supply efficiency of 92% can be achieved concurrently with superior constant current characteristics.

5. Conclusions

In this paper, a high-performance semiconductor laser power supply system is designed. By establishing the mathematical model of the LCC resonant converter, the parameters of the LCC resonant device are designed, the disadvantages of traditional control strategy algorithm are analyzed, and the inverter phase-shift compound control strategy and LADRC algorithm are proposed, which not only improves the power efficiency of semiconductor laser, but also realizes the ZVS on-switching of switching tube. It reduces the on-off loss, reduces the overall power consumption by about 3% compared with the same type of power supply, and also has a strong anti-interference ability, fundamentally avoids the limitations of PID control algorithm applied in the control of the LCC resonant converter, reduces the current and voltage stress of the LCC resonant converter components, and improves the life and reliability of semiconductor laser drive power supply. The above characteristics show the unique characteristics of the LCC resonant converter application, which is suitable for high performance semiconductor laser drive power supply.