Design of Broadband Low-RCS Array Antennas Based on Characteristic Mode Cancellation

Abstract

In this letter, a design method for low radar cross section (RCS) array antennas based on characteristic mode cancellation (CMC) is presented. Based on the characteristic mode theory (CMT), two novel microstrip elements are designed by introducing rectangular slots and cross slots, which produce 180° scattering phase difference by adjusting the size of slots. The dominant characteristic modes of the two elements achieve broadband dual-linear polarization CMC and similar radiation performances. The 4 × 4 array antenna consisting of these two antenna elements is designed. The operating band is from 4.55 GHz to 5.49 GHz (relative bandwidth 18.7%). The gain loss of the proposed array is about 0.1 dB compared to the reference array. The monostatic RCS is reduced for dual−linear polarized waves, and the 6 dB radar cross section reduction (RCSR) bandwidths are 62.3% and 35.7%, respectively. The prototype is fabricated and measured. The measured results of radiation pattern and RCS are in good agreement with the simulated results.

1. Introduction

The stealth technology of antennas has attracted much attention, and many efforts have been made to reduce the radar cross section (RCS) of array antennas in recent years. Traditional methods of antenna radar cross section reduction (RCSR) include shape modification [1], frequency selective surface (FSS) [2,3,4], radar absorbing material (RAM) [5,6,7,8], and scattering cancellation method [9,10,11,12,13,14,15,16,17,18].

Shape modification, such as loading slots on the antenna patch, can realize broadband and wide−angle RCSR. Shape modification usually requires abundant design experience, coupled with lots of optimization processes. Shape modification is mostly used in the design of element antennas and is difficult to directly apply to RCSR of array antennas. As a kind of spatial filter, FSS is transparent to in-band electromagnetic waves and has a cutoff effect on out-of-band electromagnetic waves. Therefore, a stealth radome based on FSS can reduce the out−of−band RCS of array antennas, but then it is difficult to reduce the in-band RCS. RAM can significantly reduce the RCS of array antennas over broadband, but it usually results in gain loss. It is also difficult to reduce the in-band RCS of antenna. The scattering cancellation method is based on the scattering phase difference of two objects, which can realize broadband RCSR with little gain loss. When the scattering phase difference is 180, scattering cancellation can be realized. Some researchers have focused on metasurface technology to realize scattering cancellation with antennas. In 2017, the quasi-fractal AMC structure was proposed, with a value of reflection phase difference between AMC and PEC of 180° ± 37° from 6.4 GHz to 21.7 GHz [15]. Compared to the reference antenna, the RCS of the proposed antenna was reduced from 6.0 GHz to 30.0 GHz. In 2021, three different AMC structures were proposed as the ground of a 1-D phased array [14]. Compared to the reference array antenna, the RCS was reduced from 12.0 GHz to 16.0 GHz for TE polarization and from 10.8 GHz to 15.2 GHz for TM polarization, respectively. Antenna and metasurface scattering cancellation is mainly used for RCSR of element antennas, and would increase the transverse size of the antenna element. When these structures are applied to array antennas, it will result in a scanning range that is limited due to the increase in element spacing. Some researchers propose to realize scattering cancellation by two kinds of element antennas. The element scattering cancellation of an array antenna can reduce its RCS without changing the array size, which has great application value. In 2019, a novel element with a U−shaped slot was designed [17]. The novel element had opposite scattering phases with a traditional microstrip element. The RCS of the proposed array was reduced from 4 GHz to 8 GHz, with a maximum reduction value of 23 dB. In 2022, two antenna elements were proposed by introducing L-feeding patches along the orthogonal direction to the metasurface [16]. The 6 dB RCSR bandwidth of the proposed array was from 5.1 GHz to 6.9 GHz. However, due to a lack of theoretical analysis methods, the design of the scattering cancellation elements relied more on engineering experience and optimization.

Characteristic mode theory (CMT) defines a series of orthogonal characteristic currents for electromagnetic objects of arbitrary shape. These characteristic currents are their inherent properties and can essentially explain radiation characteristics and scattering characteristics. Therefore, CMT is widely used in antenna design [19,20,21]. Recently, CMT has been used to distinguish between radiation modes and scattering modes, which are applied to realize radar cross section reduction (RCSR) of antennas [22,23,24,25,26,27,28]. In 2018, CMT was proposed to reduce the RCS of slot microstrip antennas, and RCS reduction of element antennas was realized from 2 GHz to 4 GHz [29]. The RCSR methods based on CMT have been proved to be useful for antenna element design. In 2019, metasurface design based on CMT was studied, and characteristic mode cancellation (CMC) method was proposed and applied to metasurface design. The metasurface was applied among the element antenna and the RCS reduction of the element antennas was realized from 6 GHz to 18 GHz [30]. In the above work, CMT was applied to the design of the microstrip element antenna or metasurface, and the RCS reduction of element antennas was realized. It is understood that the application of CMT to RCSR of array antennas has not been reported in public.

In this work, the scattering theory of array antennas based on CMT is studied. The RCSR method of array antennas based on CMC is proposed. For array antennas composed of two kinds of element, theoretical derivation shows that the RCS of the array antenna can be reduced by using CMC. The radiation and scattering characteristic modes of the antenna elements with rectangular slots and cross slots are proposed and analyzed. The scattering mode phase can be controlled by adjusting the size of the slots, and the broadband effective phase difference is realized. By using CMC, the array antenna is designed to realize RCSR over a wide frequency band in the case of small gain loss and broadband radiation performance. More clearly, a theory guide for scattering cancellation array antenna design is provided by CMT, which can be used for wide-band low-RCS array antenna design.

Key contributions of this paper are shown as follows:

• A method of low-RCS array antenna design based on CMC is presented.
• A dual-linear polarization low−RCS microstrip array antenna with a 6 dB RCSR bandwidth of 62.3% and 35.7% is obtained.

2. Characteristic Mode Theory and its Application to Array Antennas

Both the radiation and scattering performances of an antenna can be investigated by CMT. Therefore, CMT is not only widely applied in the radiation performance design of antenna, but is also suitable for antenna RCSR design. According to CMT, the scattering current of an object can be expressed as a superposition of a series of orthogonal and complete characteristic currents [31]:

$$\overrightarrow{J}=\sum _{n=1}^N{\alpha }_n{\overrightarrow{J}}_n={\alpha }_1{\overrightarrow{J}}_1+{\alpha }_2{\overrightarrow{J}}_2+\dots +{\alpha }_N{\overrightarrow{J}}_N$$

(1)

where ${\alpha }_n$ is the mode weighting coefficient (MWC) and ${\overrightarrow{J}}_n$ is the characteristic current of the nth mode. Furthermore, ${\alpha }_n$ denotes ${\alpha }_n=\left|{\alpha }_n\right|e^{j{\phi }_n}$ with the corresponding magnitude of ${\alpha }_n$ and phase of ${\phi }_n$.

The scattering field is determined by superposition of the characteristic fields. The scattering field of an object can be calculated as [32]:

$${\overrightarrow{E}}_S=\sum _{n=1}^N{\alpha }_n{\overrightarrow{E}}_n={\alpha }_1{\overrightarrow{E}}_1+{\alpha }_2{\overrightarrow{E}}_2+\dots +{\alpha }_N{\overrightarrow{E}}_N$$

(2)

where ${\overrightarrow{E}}_S$ is the total scattering field of the object. The scattering field for the nth mode ${\overrightarrow{E}}_n$ denotes ${\overrightarrow{E}}_n={\left|E_n\right|e}^{j{\mathsf{\gamma }}_n}$ with the corresponding magnitude of $\left|E_n\right|$ and phase of ${\mathsf{\gamma }}_n$. Therefore, ${\overrightarrow{E}}_S$ can be written as:

$${\overrightarrow{E}}_S=\sum _{n=1}^N\left|{\alpha }_n\right|{·\left|E_n\right|e}^{j({{\mathsf{\gamma }}_n+\phi }_{n)}}=\sum _{n=1}^N{A}_n{·e}^{j{\mathsf{\varphi }}_n}$$

(3)

where $A_n$ ($A_n=\left|{\alpha }_n\right|·\left|E_n\right|)$ and ${\mathsf{\varphi }}_n$ (${\mathsf{\varphi }}_n={\mathsf{\gamma }}_n+{\mathsf{\phi }}_n)$ are the amplitude and phase of the nth characteristic mode field.

The array antenna composed of two types of elements is shown in Figure 1. The scattering field of an array antenna can be approximately expressed as the product of the element scattering field and the scattering array factors when the mutual influence among the antenna elements is ignored. The scattering fields of an antenna element can be expressed as in Equations (2) and (3). The scattering fields of the two subarrays could then be respectively written as:

$${\overrightarrow{E}}_S^1=\left(\sum _{n=1}^N{\alpha }_n·{\overrightarrow{E}}_n^1\right)·{\sigma }^1$$

(4)

$${\overrightarrow{E}}_S^2=\left(\sum _{m=1}^M{\beta }_m·{\overrightarrow{E}}_m^2\right)·{\sigma }^2$$

(5)

where ${\alpha }_n$ and ${\beta }_m$ are the MWC of the two elements, and ${\sigma }^1$ and ${\sigma }^2$ are the scattering array factors of the two subarrays. When these two kinds of element have the same quantity and same array form, the scattering array factors are the same (${\sigma }^1={\sigma }^2={\sigma }_0$). Therefore, the total scattering field of the array antenna composed of these two subarrays with element 1 and element 2 could be given as:

$$\begin{gathered} {{\overrightarrow{E}}_S=\overrightarrow{E}}_S^1{+\overrightarrow{E}}_S^2=\left(\sum _{n=1}^N{\alpha }_n·{\overrightarrow{E}}_n^1+\sum _{m=1}^M{\beta }_m·{\overrightarrow{E}}_m^2\right)·{\sigma }_0 \\ =\left(\sum _{n=1}^N{A}_n{·e}^{j{\varphi }_n}+\sum _{m=1}^M{B}_m{·e}^{j{\psi }_m}\right)·{\sigma }_0 \end{gathered}$$

(6)

where $A_n$ and ${\mathsf{\varphi }}_n$ are the amplitude and phase of the characteristic mode field of element 1, and $B_m$ and ${\psi }_m$ are the amplitude and phase of the characteristic mode field of element 2. It can be concluded from Equation (6) that the scattering field of the array antenna can be reduced by decreasing the superposition of the scattering characteristic fields of these two elements. When the scattering amplitude of the two elements has equal value ($A_n=B_m=A_0$), Equation (6) can be written as:

$${\overrightarrow{E}}_S=A_0·\left(\sum _{n=1}^N{e}^{j{\varphi }_n}+\sum _{m=1}^M{e}^{j{\psi }_m}\right)·{\sigma }_0$$

(7)

For example, when the two element antennas have two dominant modes, the amplitude of the scattering field of the array can be calculated as:

$$\begin{gathered} {\overrightarrow{E}}_S=A_0·\left[\left(e^{j{\varphi }_1}+e^{j{\varphi }_2}\right)+\left(e^{j{\psi }_1}+e^{j{\psi }_2}\right)\right]·{\sigma }_0 \\ =A_0·\left[\left(e^{j{\varphi }_1}+e^{j{\psi }_1}\right)+\left(e^{j{\varphi }_2}+e^{j{\psi }_2}\right)\right]·{\sigma }_0 \end{gathered}$$

(8)

In this paper, the phase of the characteristic mode is regulated by the modification technology. When ${\varphi }_1-{\psi }_1=180°$ or ${\varphi }_2-{\psi }_2=180°$, the CMC can be realized and the RCS is reduced. When the scattering phase difference of the characteristic mode field of two elements is 180°, ${\overrightarrow{E}}_S=0$ will be obtained to achieve minimal RCS. Therefore, the RCS of an array antenna can be reduced by introducing CMC technology.

3. Design of the Antenna Element Based on CMT

In this paper, two novel antenna elements are proposed based on CMT. The phase of the characteristic mode of an element is regulated by the shape modification technology to realize CMC. The ground plane would be connected when the patch elements are arranged in an array. Therefore, in order to maintain the mode phase characteristics of elements in the array, the modification is carried out on the radiation patch. Firstly, a novel antenna element with rectangular slots (called antenna A) is designed to achieve CMC, and the reference antenna is a traditional square patch antenna element. The radiation characteristic currents of the reference antenna and antenna A are similar, and the scattering characteristic currents are opposite. Next, in order to expand the CMC bandwidth, another novel antenna element (called antenna B) etched with cross slots is designed. Antenna A and antenna B can realize broadband dual-polarization CMC, which lays a foundation for the wide-band RCS reduction for the array antenna.

3.1. Characteristic Mode Analysis of the Reference Antenna and Antenna A

The geometries of reference antenna and antenna A are shown in Figure 2 and their main dimension parameters are shown in

Table 1. Parameters of the design elements (Unit: mm).

Symbol	Value	Symbol	Value	Symbol	Value	Symbol	Value
L	30	wsb	0.2	lsa	18	da	4
t	6	wdb	1	wda	5.5	lr	14.5
lb	19	ldb	9	la	15.5	ha	3.5
lsb	20	db	8	lda	9	lsr	21

. The reference antenna and antenna A have similar structures and the dielectric material is polytetrafluoron (${\epsilon }_r=2.2$, $\tan {\delta }_r=0.009$). The lower layer is a ground plane structure with rectangular slots which is used to adjust the antenna resonant frequency. The middle layer adopts an L-shaped coupling feed structure to widen the operating bandwidth. For the reference antenna, the upper layer is a square patch, as shown in Figure 2a. For antenna A, the upper patch is a square patch with two rectangular slots through which the characteristic mode can be regulated, as shown in Figure 2b. For a y-polarized wave, the equivalent electrical length of the scattering current can be increased, which impacts the phase of the characteristic mode.

The simulation results of the reflection coefficient (S₁₁) and far-field patterns of the two antenna elements are shown in Figure 3. The two elements have the same resonant frequency and similar far−field patterns in both the xoz plane and yoz plane at 5.1 GHz. The operating frequency with the reflection coefficient less than −10 dB is from 4.64 GHz to 5.52 GHz.

To maintain stability while effectively reducing RCS, the radiation and scattering characteristic modes analyses need to be undertaken.

The radiation characteristic modes analysis is firstly carried out. The MWC (${\alpha }_n$) and characteristic electric field (${\overrightarrow{E}}_n$) are analyzed. The value of $A_n$ can be calculated, as shown in Figure 4. Mode 1 is the dominant mode for the two antenna elements. The characteristic current distribution of the two dominant modes is similar, as shown in Figure 5. Therefore, the radiation properties of these two antennas are similar.

The scattering characteristic mode analyses of the reference antenna and antenna A are performed from 4.0 GHz to 8.0 GHz. For the y−polarized incident wave, the MWC (${\alpha }_n$) and characteristic electric field (${\overrightarrow{E}}_n$) of the two elements are simulated. The amplitudes ($A_n$) of the first six modes are calculated in Figure 6 and the phases (${\mathsf{\varphi }}_n$) of the two dominant modes are given Figure 7. It can be seen from Figure 6 that modes 1 and 2 are the dominant modes for the reference antenna and antenna A. The rectangular slots on antenna A lengthen the current path of mode 1 and mode 2, so $A_n$ of the modes moved to lower frequency. Figure 7a shows ${\mathsf{\varphi }}_n$ of the reference antenna and antenna A. The value of the phase difference is shown in Figure 7b, in which ${\Delta \mathsf{\varphi }}_1$ is the phase difference between mode 1 of the reference antenna and antenna A and ${\Delta \mathsf{\varphi }}_2$ is the phase difference between mode 2 of the reference antenna and antenna A. For out-band 4.4 GHz and in-band 5.0 GHz, the values of $A_n$ of the cancelled modes (mode 1 and mode 2) are similar. The characteristic current and far−field of the two cancelled modes are shown in Figure 8 and Figure 9. The characteristic current of the reference antenna is opposite to that of antenna A, and the characteristic far−field patterns of the two modes are similar. Therefore, mode 1 of the reference antenna and antenna A form cancellation at 4.4 GHz. Mode 2 of the reference antenna and antenna A form cancellation at 5.0 GHz.

3.2. Characteristic Mode Analysis of Antenna B

Though the CMC between the reference antenna and antenna A is realized at 4.4 GHz and 5.0 GHz, the mode phases no longer satisfy CMC conditions at higher frequencies. To achieve wideband RCSR, the characteristic modes of the reference antenna should be changed. Therefore, a pair of cross slots is etched on the radiation patch (called antenna B) to control the scattering phase, as shown in Figure 10. The antenna structure is similar to antenna A and the main dimension parameters are shown in Table 1. The cross slots can expand the CMC bandwidth and realize dual−polarization scattering cancellation.

The radiation characteristic modes of antenna B are first analyzed. Figure 11 shows $A_n$ and the characteristic current distribution. It can be seen from Figure 11a that mode 1 is the dominant mode of antenna B. The characteristic current distribution of mode 1 of antenna B is similar to antenna A, as shown in Figure 11b.

The simulation results of the reflection coefficient (S11) and far-field patterns of the two antenna elements are shown in Figure 12. It can be seen that antenna A and antenna B have the same resonant frequency and similar far-field patterns in both the xoz plane and yoz plane at 5.1 GHz. The operating frequency with a reflection coefficient less than −10 dB is from 4.55 GHz to 5.49 GHz.

The scattering characteristic mode analysis of antenna B is performed from 4.0 GHz to 8.0 GHz. For the y−polarized incident wave, the MWC (${\alpha }_n$) and characteristic electric field (${\overrightarrow{E}}_n$) of antenna B are analyzed. The key parameters ($A_n$ and ${\mathsf{\varphi }}_n$) of the characteristic modes are calculated as shown in Figure 13. It can be seen from Figure 13a that modes 1 and 2 are the dominant modes. The phase (${\mathsf{\varphi }}_n$) of the dominant scattering modes is calculated, as shown in Figure 13b. The phase difference between antenna A and antenna B is shown in Figure 13c. There, ${\Delta \mathsf{\varphi }}_3$ is the phase difference between mode 1 of antenna A and antenna B and ${\Delta \mathsf{\varphi }}_4$ is the difference between mode 2 of antenna A and antenna B, and ${\Delta \mathsf{\varphi }}_4$ is about 180° from 5.6 GHz to 7.6 GHz. The values of $A_n$ of the cancelled modes are similar. Therefore, broadband scattering cancellation is realized. The characteristic currents and far-field patterns of antenna A and antenna B at 7.0 GHz are shown in Figure 14. The current direction of mode 2 of antenna A is opposite to that of mode 2 of antenna B, and the far-field patterns of the modes are similar. Therefore, a cancellation effect of antenna A and antenna B is obvious, and the RCSR is expected to achieve between 4.0 GHz and 8.0 GHz for the y-polarized wave.

For the x-polarized incident wave, the scattering characteristic mode analysis of antenna A and antenna B is performed from 4.0 GHz to 8.0 GHz. Figure 15 and Figure 16 show the calculations of $A_n$ and ${\mathsf{\varphi }}_n$ of the scattering characteristic modes. It can be seen from Figure 15 that modes 1 and 2 are the dominant modes for antenna A and antenna B. The mode amplitude ($A_n$) is small in the antenna operating band, which originates from co-polarization absorption due to antenna impedance matching. The scattering phase (${\mathsf{\varphi }}_n$) is shown in Figure 16a and the phase difference between antenna A and antenna B is shown in Figure 16b, where ${\Delta \mathsf{\varphi }}_5$ is the phase difference between mode 1 of antenna A and antenna B, and ${\Delta \mathsf{\varphi }}_6$ is the phase difference between mode 2 of antenna A and antenna B. There, ${\Delta \mathsf{\varphi }}_6$ is 175° at 6.0 GHz and the corresponding values of $A_n$ are about 0.015. Therefore, a cancellation effect of antenna A and antenna B is produced outside of operating band. It can be concluded that the RCSR of an array antenna composed of antenna A and antenna B can be obtained from 4.2 GHz to 6.6 GHz for the x-polarized incident wave.

4. Design of the Array Antennas

The 4 × 4 array antenna consisting of antenna A and antenna B is designed and the number of antenna As and antenna Bs is equal. Considering the influence of scattering coupling on RCS of array antennas, the RCS of different array forms is studied. The array antennas are in forms of ABAB, ABBA, AABB, and checkerboard respectively, as shown in Figure 17. The size of the four arrays is 120 mm × 120 mm. The RCS simulation results of the four arrays are shown in Figure 18. As seen from the Figure 18, the RCS reduction of different array forms is all realized in the 4 GHz–8 GHz band, but the RCS of the ABAB array forms is the lowest. Therefore, the form ABAB is used to conform to CMC array antennas. The element structures are the same in the x−direction and are arranged in the form of ABAB in the y−direction.

A conventional 4 × 4 patch array antennas consisting of the reference antenna is shown in Figure 19. The size of the reference array is 120 mm × 120 mm. The radiation far−field pattern simulation results of the array antennas are shown in Figure 20. The far−field patterns of the reference array and proposed array are basically the same and the gain difference is about 0.1 dB.

The simulation results comparison of the monostatic RCS between the two arrays is shown in Figure 21. For the x-polarized incident wave, the RCS of the proposed array is reduced from 4.0 GHz to 8.0 GHz compared with the reference array. The maximum value of RCSR is 32.3 dB at 6 GHz and the average value of in−band RCSR is 9.5 dB. The 6 dB RCSR bandwidth of the proposed array antenna is from 4.6 GHz to 6.6 GHz. Combined with the scattering characteristic mode analysis for the x-polarized incident wave, it can be concluded that the in-band RCSR of the x−polarized incident wave comes from the absorption and the out-of-band RCSR comes from CMC of the two antenna elements. For the y-polarized incident wave, the RCS of the proposed array is reduced from 4.0 GHz to 8.0 GHz compared with the reference array. The maximum reduction is 24 dB at 5.6 GHz, and the average RCSR value is 13.1 dB. The average value of in−band RCSR is 16 dB. The 6 dB RCSR bandwidth of the proposed antenna array is from 4.2 GHz to 8.0 GHz. Combining the scattering characteristic mode analysis for the x-polarized incident wave, it can be concluded that the RCSR of the y-polarized incident wave comes from CMC of the two antenna elements.

The comparison of the radiation and RCS characteristics among typical low-RCS array antennas and the proposed array antenna is shown in

Table 2. Comparison with typical low-RCS arrays.

Ref.	$\mathbf{Electric}\mathbf{at}\mathbf{Size}\left({\mathit{\lambda }}^3\right)$	OB (GHz)	Gain (dB)/Aperture Efficiency	Co/Cross−Polarization RCSR Mean Value in OB (dB)	Co/Cross−Polarization RCSR B.W. (GHz)		Method
Ref. [4]	2.2 × 2.2 × 0.8	-	15.8/62.5%	11.8/19.8	-	6–18 (100%)	FSS
Ref. [8]	4.7 × 4.7 × 0.7	26.7–34.2 (24.6%)	20/36.7%	12/12	-	28–48 (52%)	Absorptive
Ref. [16]	1.8 × 1.8 × 0.05	5.05–5.42 (7.1%)	-	-/16.5	-	4–8 (66.7%)	A.E.C.
Ref. [17]	2 × 2 × 0.1	4.75–5.25 (10%)	-	13/13	4–8 (66.7%)	4–8 (66.7%)	A.E.C.
This work	2 × 2 × 0.1	4.55–5.49 (18.7%)	16.6/89%	14.7/22.8	4–8 (66.7%)	4–8 (66.7%)	CMC

OB: operating bandwidth. B.W.: bandwidth. A.E.C.: antenna element cancellation.

. The methods of FSS, absorbing material loading technology, and cancellation technology are listed. For antenna size, the transverse size and profile height of the antenna array proposed in this work are nearly the same as the size of the array antenna in Refs. [16,17], and the profile height is reduced compared with array antenna in Refs. [4,8]. For radiation performance, the operating bandwidth of the array antenna in this work achieves significantly wider operating bandwidth and it is only slightly lower than the array design based on FSS in Ref. [8]. The aperture efficiency of the proposed antenna is higher. For scattering performance, the cross−polarization RCSR bandwidth is nearly the same as other array antennas. The mean value of RCSR in operating bandwidth is improved for both co-polarization and cross−polarization. Compared with the low−RCS array antennas designed by antenna element cancellation (A.E.C.), the operating bandwidth is wider and the RCSR mean value in the operating bandwidth is bigger.

5. Experimental Results and Discussion

The prototype consists of an antenna radiation array and power dividers. The beam synthesis for the 16 units of the array is realized by power dividers. The antenna radiation array is composed of two layers of PCB structure. These two layers are fabricated by printed board (PCB) technology. The upper PCB is printed with the radiation patch of the antenna. Both sides of the lower PCB are printed with a different structure. The upper surface is printed with the antenna feeder, and the lower surface is printed with rectangular slots. Finally, the two layers of PCB structure are spliced together. The prototype of the proposed array antenna is shown in Figure 22a.

The S₁₁ of the antenna prototype is measured by a vector network analyzer. Antenna elements are fed by coaxial probes. The measured and simulated S₁₁ of antenna A and antenna B are shown in Figure 23a. The measured reflection coefficient less than −10 dB is from 4.54 GHz to 5.46 GHz, which is almost the same as the simulated results.

The realized gain and radiation patterns measurement of the antenna prototype are carried out in the microwave anechoic chamber, as shown in Figure 22b. The far−field condition is satisfied between the antenna prototype and the standard horn antenna. The array antenna is fed by a power divider when measuring its far-field patterns. The far−field patterns of the fabricated array antenna are tested, as shown in Figure 23b. Because some energy is lost in the SMA connector and coaxial line, the measured gain is 0.5 dB lower than the simulated result.

The scattering performance of the array antenna prototype is tested in the compact range anechoic chamber as shown in Figure 24. Array antenna elements are all matched 50 Ω loads, and the prototype is placed on a low RCS foam support. The plane wave of the test field is vertically incident on the antenna radiation surface, and the frequency sweep test is carried out. The measured and simulated monostatic RCS results of the proposed antenna for vertical incident plane waves with x−polarization and y−polarization are shown in Figure 25. The measured results are in good agreement with the simulated results. The difference between the measured results and simulated results is mainly due to assembly error and measurement error.

In order to illustrate the measured results, the impact of the manufacturing error is analyzed. The dielectric constant of the PCB printed board is more accurate. The pattern-etching machining technology is mature and the etching precision is higher. Therefore, the machining error introduced by the PCB can be ignored. The gap and the displacement between the double PCB layers play a major role in manufacturing error of the prototype. For the y−polarized incident wave, we calculated RCS in conditions of the space gap widths of 0 mm, 0.3 mm, and 0.5 mm respectively, as shown in Figure 26a. We also calculated RCS in conditions of the displacement sizes of 1 mm, 2 mm, and 3 mm respectively, as shown in Figure 26b. It can be found that the RCS simulation results with the error show the same trend of change as the measured results. Therefore, the gap and the displacement tolerances should be strictly controlled in the manufacturing process.

6. Conclusions

In this work, a broadband RCSR designed method of array antenna based on CMC is presented and experimentally verified. Through loading rectangular slots and cross slots on the patch, the phase (${\mathsf{\varphi }}_n$) of the characteristic modes can be controlled. When the scattering phase difference between the dominant characteristic modes of the two elements achieves 180°, characteristic mode cancellation is produced. The two proposed antenna elements, which have similar radiation performances and broadband dual−linear polarization scattering cancellation, are used to form array antenna. Simulated and measured results show that the operating frequency with a reflection coefficient less than −10 dB of the two elements is from 4.55 GHz to 5.49 GHz and the gain loss of the proposed array is less than 0.1 dB. The monostatic RCS is reduced for dual−linear polarized waves, and the 6 dB RCSR bandwidth is 62.3% and 35.7%, respectively.

Therefore, characteristic mode cancellation provides a new RCSR solution for array antennas. Furthermore, this method is not limited to the form of antenna and can be applied to designs of various types of array antenna. In the design of low−RCS array antenna RCSR based on CMC, it relies on the characteristic mode analysis of the element, rather than the analysis of the whole array. Therefore, the RCS of the array antenna will be affected by the coupling among the antenna elements of the array, which will be further studied.