A Study of 500 W/250 mm Inductive Power Transfer System for Television Appliance

Abstract

This study presents the design, analysis, and experimental validation of a 500 W inductive power transfer (IPT) system with a transmission distance of 250 mm for television applications. The proposed system incorporates an innovative wireless pad design featuring a four-teeth magnetic structure and an LCC-S compensation topology to optimize coupling coefficients, reduce copper losses, and improve overall efficiency. The system’s robustness under misalignment and load fluctuations was validated, with experimental results confirming over 80% efficiency for optimal configurations. The findings also highlight the sensitivity of the system to switching frequency variations, emphasizing the need to maintain resonance conditions for maximum power transfer. Compared to existing designs, the proposed system demonstrates superior performance in long-distance wireless power transfer, making it a promising solution for high-power applications in home appliances.

1. Introduction

Wireless power transfer (WPT) systems are widely utilized in various applications due to their convenience. Mobile phones and electric vehicles have already been commercialized, and research in these areas is actively ongoing. In the field of home appliances, products utilizing wireless power transfer technology are being developed for convenience and esthetic reasons [1,2,3].

Primarily, research on wireless power transfer technology for home appliances applies inductive power transfer (IPT) systems, where power transmission distances are mostly below 10 mm, as in electric toothbrushes, kettles, and cooktops. Various studies are being conducted on the practical implementation of complete wireless power transfer technology over distances exceeding 100 mm. For high-end home appliances, the power consumption is significant due to their high performance, making the application of wireless power transmission technology for hundreds of millimeters and hundreds of watts crucial.

Research on wireless power transfer (WPT) for a distance of 200 mm using an intermediate coil was conducted in [4]. Although it achieved a maximum power transfer of 150 W with an efficiency of 80%, the large size of the intermediate coil (1490 mm × 984 mm) introduced significant drawbacks in terms of system volume and cost. Furthermore, the lack of flux shielding made practical application challenging. In [5], a planar coil winding and a teeth structure were proposed to improve misalignment characteristics, achieving 200 mm WPT. Aluminum shielding was used to enable flux containment, but the received power was only 76.8 W, and the wireless pad size (510 mm × 300 mm × 45 mm) remained relatively large. In [6], a long-distance WPT of 5000 mm was demonstrated using a dipole structure. However, the core length of 3000 mm and the absence of flux shielding made practical application difficult. In [7], power transmission was achieved over a distance of 300 mm, but only 1 W of power was transferred, limiting its practicality for real-world applications. In a study by [8], they proposed a dipole-type WPT system capable of wirelessly transferring power over 1000 mm. However, the transmitter relied on a magnetic material measuring 2500 mm × 500 mm × 100 mm, leading to challenges in practical implementation. Additionally, it achieved only 150 W of power transfer and exhibited significant performance degradation due to misalignment. Partial shielding was applied only to the receiver, limiting real-world applicability. In [9], planar coil structures achieved 500 mm WPT, but with an output power of only 1.5 W and a low efficiency of 70%, rendering it impractical. In [10], multilayer relay structures were implemented to develop IPT technology for refrigerators, supplying power to fans, LEDs, and cameras. Despite achieving a transmission distance of 150 mm, the low output voltage and total power (below 10 W) posed limitations. The system also required a high switching frequency of 6.78 MHz, complicating practical deployment. In [11], 100 W was transferred over 300 mm, but as described in [3], the lack of shielding considerations hindered practical implementation and scalability for higher power levels. In [12], single-tube loop coil and multi-turn copper wire coil technologies were used to transfer power over 300 mm, but the low output power of 1 W and lack of shielding again rendered it unsuitable for practical applications. Research presented in [13] examined WPT systems for home appliances using multiple coils to increase coupling coefficients. Although 100 W was transferred over 300 mm, the efficiency was below 80%, and the absence of flux shielding limited its practical applicability. In [14], an omnidirectional WPT system for home appliances was proposed, demonstrating power transfer across various angles. However, due to the omnidirectional nature, the actual coil spacing resulted in low efficiency and power transfer below 100 W. In [15], coupling coefficient optimization was studied, but the transmission distance was limited to 40 mm, with a maximum transfer power of only 66 W. The lack of flux shielding further limited its practicality. In [16], a dipole coil system was constructed for 400 mm long-distance WPT. Although long-distance power transfer was achieved, the efficiency at maximum distance was only 8.8%. Additionally, the wireless pad measured 800 mm in length, and the absence of flux shielding made practical implementation challenging. At CES 2021, a system capable of wirelessly powering a 40-inch TV was unveiled. While achieving a maximum transmission distance of 500 mm, the large size of the transmitter and receiver coils (40 inches) and the limited power capacity of 120 W made it unsuitable for larger TVs [17]. In [18], a 250 mm power transmission was achieved using a structure with six intermediate coils, fundamentally limiting practicality due to the inherent drawback of having additional coils between the transmitter and receiver, and flux shielding was also not considered. In [19], planar coils were studied for 700 mm long-distance power transfer. However, the efficiency was only 28%, and the transmitted power was limited to 2 W, both of which were impractically low. Flux shielding was also not considered.

Despite numerous studies on long-distance WPT, most suffer from reduced output power and efficiency at extended distances. To overcome these limitations, larger wireless pads are often used. As shown in

Table 1. Comparison of the previous research of a long-distance WPT system.

Ref.	Air Gap [mm]	Power [W]	Coil Structure	Efficiency [%]	Shielding
[4]	200	150	Intermediated planar	80	X
[5]	200	76.8	Planar, Teeth	86.9	O
[6]	5000	5	Dipole	6	X
[7]	300	0.97	Planar	15.04	X
[8]	215	1000	Dipole	71	Δ
[9]	500	1.5	Planar	70	X
[10]	150	10	Planar	85	X
[11]	200	320	Planar	88	X
[12]	300	0.908	Planar	1.68	X
[13]	300	100	Planar	80	X
[14]	200	100	Rotary planar	49.8	X
[15]	40	66	Planar	62	X
[16]	400	200	Dipole	8.8	X
[17]	500	120	Planar	-	X
[18]	250	12.5	Intermediated planar	68.6	X
[19]	290	22	Planar	6	X

, most studies did not consider flux shielding, and the addition of shielding is expected to reduce efficiency and output power, making practical application difficult.

This paper proposes a WPT system for TVs capable of transferring 500 W over a distance of 250 mm with aluminum shielding. To achieve long-distance power transfer, an optimal design of the wireless pad, a critical component of WPT, was performed. The design process included analyzing coil shapes for long-distance power transfer, optimizing magnetic paths with magnetic teeth to increase the coupling coefficient, and varying the number of turns to optimize copper losses in the transmitter and receiver pads. Additionally, misalignment was considered and verified through analysis and experiments. The performance improvements over previous studies were demonstrated using a 500 W testbed.

2. Analysis of the Long-Distance Wireless Power Transfer System for TVs

The wireless power transfer method used in this paper is illustrated in Figure 1, consisting of a DC-AC inverter, primary resonant network, primary wireless pad, secondary wireless pad, AC-DC rectifier, and final DC-DC converter. The resonant network of the transmitter and receiver should use a higher-order topology considering the low mutual inductance M due to the low coupling coefficient k. Given the load characteristics of the TV, an LCC-S topology was chosen for its constant voltage output characteristics.

2.1. Wireless Pad Design Consideration

The wireless charging system for TVs aims to deliver 500 W of power over a physical distance of 250 mm between the transmitter and receiver. The design of the wireless pad is critical for transmitting higher power over a longer distance compared to existing home appliance wireless power transfer systems. The wireless pad must incorporate aluminum shielding to prevent magnetic flux leakage and utilize magnetic materials to increase inductance.

Different coil shapes are possible depending on the application. Typical shapes for IPT coils include circular and dipole configurations, both of which are analyzed in this study. The dimensions for the wireless pad used in this study were determined to be 300 mm × 100 mm × 20 mm to ensure practical product application. Considering the high switching frequency of 100 kHz, a 0.1 mm diameter Litz wire with 1150 strands was used for the coils. To prevent magnetic flux leakage, aluminum shielding was added to both the transmitter and receiver pads with a 15 mm gap between them.

Table 2. System key parameters.

Parameter	Symbol	Value
Input voltage	UIN	380 V
Rectifier output voltage	UDC	100 V
Output power	PDC	500 W
Switching frequency	FSW	100 kHz
Wireless pad distance	-	250 mm

shows summarized system parameters, and basic wireless pad structure of this paper is shown in Figure 2.

2.2. Resonant Network Configuration

The structure of the IPT system with an LCC-S resonant network can be equivalently represented using a loosely coupled transformer and an AC output U_ab derived from the AC input voltage source U_AB generated by the inverter, as shown in Figure 3. Calculating the coil current size is crucial for designing the wireless pad’s coil. At the resonant frequency, the transmitter coil current I_P is calculated as follows:

$$I_{\mathrm{P}}={\displaystyle \frac{U_{\mathrm{AB}}}{j\omega L_{\mathrm{IN}}}}$$

(1)

where L_IN is the input resonant inductor and ω is the angular frequency. The transmitter coil current I_S is equal to the output AC current I_ab, which can be computed using the output AC voltage U_ab and the equivalent output AC resistance R_o.ac.

$$I_{\mathrm{S}}=I_{\mathrm{ab}}={\displaystyle \frac{U_{\mathrm{ab}}}{R_{\mathrm{o}.\mathrm{ac}}}}$$

(2)

The output AC voltage U_ab is calculated as follows:

$$U_{\mathrm{ab}}={\displaystyle \frac{M\cdot U_{\mathrm{AB}}}{L_{\mathrm{IN}}}}$$

(3)

where M represents the mutual inductance between the transmitter and receiver coils. To obtain the desired output voltage, M must be derived, and L_in designed accordingly. In this paper, the primary-side DC-AC inverter is configured as a full-bridge structure, which produces a square wave voltage alternating between +380 V and −380 V by controlling four switches in half-duty cycles. For ease of analysis, a fundamental harmonic analysis (FHA) method is applied.

According to Equation (2), since the transmitter current is the same as the output current, the specifications in this paper may not be large, but due to the low mutual inductance from the long distance, a small Lin value is anticipated, leading to high Ip current. To mitigate this, pad structure design to increase M through the coupling coefficient is necessary. The relationship between coupling coefficient k and mutual inductance M is as follows:

$$k={\displaystyle \frac{M}{\sqrt{L_{\mathrm{P}}{L}_{\mathrm{S}}}}}$$

(4)

Using Equations (1), (3), and (4), I_P can be expressed in terms of M and k, allowing for effective system design.

$$I_{\mathrm{P}}={\displaystyle \frac{U_{\mathrm{ab}}}{j\omega M}}={\displaystyle \frac{U_{\mathrm{ab}}}{j\omega k\sqrt{L_{\mathrm{P}}{L}_{\mathrm{S}}}}}$$

(5)

According to Equation (5), methods to reduce the transmitter coil current IP in the LCC-S topology include the following: (1) increasing the coupling coefficient by improving the shape of the pad; (2) increasing the inductance of either the transmitter or receiver coils; and (3) lowering the output AC voltage U_ac to reduce IP. This can be achieved by modifying the rectifier structure at the receiver to differentiate between the output DC voltage U_DC and U_ac; by reducing U_ac, IP can be decreased without changing other parameters, as expressed in (5). This paper discusses the analysis of these three methods in the context of pad design and circuit topology.

2.3. Rectifier Configuration

The rectifier at the receiver is typically a full-bridge rectifier as shown in Figure 4a. The relationship between output voltage U_DC and U_ac in the full bridge is as follows:

$$U_{\mathrm{DC}}={\displaystyle \frac{\pi \sqrt{2}}{4}}{U}_{\mathrm{ab}}={\displaystyle \frac{\pi \sqrt{2}}{4}}{\displaystyle \frac{M{U}_{\mathrm{AB}}}{L_{\mathrm{IN}}}}$$

(6)

Figure 4b shows the voltage doubler (also called the Delon circuit). The output U_DC, when using the voltage doubler, is double that of U_ac, so the output voltage U_DC is as follows:

$$U_{\mathrm{ab}}={\displaystyle \frac{M\cdot U_{\mathrm{AB}}}{L_{\mathrm{IN}}}}{U}_{\mathrm{DC}}={\displaystyle \frac{2\pi \sqrt{2}}{4}}{U}_{\mathrm{ab}}={\displaystyle \frac{2\pi \sqrt{2}}{4}}{\displaystyle \frac{M{U}_{\mathrm{AB}}}{L_{\mathrm{IN}}}}$$

(7)

The half-wave rectifier, stacked as two units, has a greater ripple current stress on the capacitors compared to the full-bridge rectifier. However, since the output DC current in this study is relatively small at 5 A, designing the capacitors does not pose significant challenges. Consequently, with half of U_ac and fixed values for M and switching frequency, the current I_P is reduced to half that of the full-bridge rectifier. However, the current I_S doubles, which may increase the losses in the receiver coil. In IPT systems, the copper losses P_Cop.tot occurring in both the transmitter and receiver coils account for a significant portion of the total losses, calculated according to the following equation:

$$P_{\mathrm{cop}.\mathrm{tot}}=I_{\mathrm{P}}^2\cdot R_{\mathrm{P}}+I_{\mathrm{S}}^2\cdot R_{\mathrm{S}}$$

(8)

where R_P and R_S are resistance of transmitter coil and receiver coil, respectively. Therefore, it is necessary to analyze the effects of the rectifier configuration. The resonant network components, including L_IN, C_p, C_s, and C_F, are calculated as follows for each configuration [20].

$$L_{\mathrm{IN}}={\displaystyle \frac{M\cdot U_{\mathrm{AB}}}{U_{\mathrm{ab}}}},C_{\mathrm{P}}={\displaystyle \frac{1}{{\omega }^2{L}_{\mathrm{IN}}}},C_{\mathrm{S}}={\displaystyle \frac{1}{{\omega }^2{L}_{\mathrm{S}}}},C_{\mathrm{F}}={\displaystyle \frac{1}{{\omega }^2(L_{\mathrm{P}}-L_{\mathrm{IN}})}}.$$

(9)

3. Design of the Long-Distance Wireless Pad and Analysis of Copper Losses

The wireless pad is the most significant loss component in the IPT system, making its design crucial for system efficiency. For long-distance wireless power transmission, the design follows these steps:

(1)

Analyze the coupling coefficient k based on the basic wireless pad shape.

(2)

Improve the magnetic paths by adding magnetic teeth and analyzing the maximum size and variability of the coupling coefficient k.

(3)

Analyze the coil turns considering electrical characteristics.

The design prioritizes the shape of the pad and teeth structure affecting the coupling coefficient, followed by the inductance and resistance of the coils determined by the number of turns.

3.1. Performance Analysis Based on Wireless Pad Shape

The wireless pad designed for TV wireless power transmission is elongated along the horizontal axis, with coils designed in two configurations: circular and dipole. The coupling coefficient k for both configurations was compared using FEM simulations.

Figure 5 shows the results of the FEM simulations for two different coil shapes. In the case of the circular type, as shown in Figure 5a, it can be observed that the ratio of magnetic flux coupling in the transmitter is high, while the flux coupling with the receiver is low. In contrast, the dipole configuration demonstrates that the magnetic flux generated by the transmitter couples more effectively with the receiver. The simulation results confirm a significant difference in the coupling coefficients, with the dipole structure having a coupling coefficient of 0.048 and the circular pad having a coefficient of 0.007. For TVs, where positional variations are minimal once installed, the properties of the circular pad, which is robust against spacing, are not suitable. Instead, the dipole structure, which can achieve a higher coupling coefficient, is expected to reduce the transmitter pad current I_P as indicated by Equation (5). Therefore, for long-distance power transfer, the basic coil structure is more suitable as a dipole configuration.

Figure 6a,b show the structure of the basic dipole configuration with the addition of magnetic teeth to improve the magnetic path. The diameter of the coils used is 5 mm, and to accommodate the additional teeth without exceeding the maximum height, the thickness of the teeth is set at 10 mm. For practical fabrication, the widths of the teeth are set at 50 mm for three teeth and 37.5 mm for four teeth to maintain the same winding window area. FEM simulations were performed to verify the coupling coefficient and self-inductance based on the number of teeth when the coil has 20 turns. Comparing the performance of three and four teeth, the self-inductance with three teeth is 91.16 μH, and the coupling coefficient is 0.047. In the case of four teeth, the self-inductance is 92.85 μH, and the coupling coefficient is 0.049. The four teeth structure has a greater magnetic flux path between the transmitter and receiver, resulting in a higher coupling coefficient and also being advantageous against spacing due to the multiple teeth. Therefore, in this paper, a configuration with four teeth is adopted as the basic structure.

Next, the positions of the teeth were varied, and two configurations were simulated as shown in Figure 7. In each case, the teeth are positioned at both ends of the pad as in Figure 7a, or the coils are designed to be positioned as in Figure 7b. Consequently, the distance between the teeth changes to 50 mm in Figure 7a and 30 mm in Figure 7b. In Figure 7a, it can be observed that the coils are more concentrated compared to Figure 7b, which has one less winding section, totaling four sections. The self-inductance and coupling coefficient for the structure in Figure 7a are the same as those in Figure 6b, at 92.85 µH and 0.049, respectively. For the structure in Figure 7b, the self-inductance is 62.85 µH and the coupling coefficient is 0.046. Therefore, the structure in Figure 7a, which demonstrates superior magnetic characteristics, is selected as the final configuration for the pad.

Misalignment occurs in wireless pads depending on their installation position. In this study, the coupling coefficients under x-axis variations were analyzed based on FEM simulation for circular type, dipole type, three-teeth, and four-teeth structures, as shown in Figure 8. In all results, the four teeth structure exhibited a higher coupling coefficient, which is expected to improve the system performance.

3.2. Analysis of Coil Turns Considering Electrical Characteristics

Finally, the performance is compared and analyzed based on the number of coil turns. As the number of coil turns increases, both the self-inductance and mutual inductance also increase. The increase in self-inductance allows for a reduction in I_P according to Equation (5). However, the increase in the number of turns can lead to higher resistance in the coils, resulting in increased copper losses. Therefore, it is essential to consider the reduction in I_P against the increase in resistance by adjusting the number of turns in both the transmitter and receiver coils. Additionally, performance analyses are conducted by varying the structures of the transmitter and receiver to evaluate the combination of dipole structures and multiple E configurations. The effects on I_P and I_S based on the receiver rectifier structure will also be considered to provide a comprehensive analysis of the characteristics based on the pad combinations.

For the coils used in the fabrication, a strand diameter of 0.1 mm and 1150 strands were utilized, with a total coil diameter of 5 mm made from Litz wire. Generally, when using multiple strands and with a large overall simulation space, issues with minimum mesh in the simulation can arise, leading to decreased accuracy when interpreting the AC resistance component through FEM simulations. Therefore, in this paper, the basic dipole structure without teeth and the multiple E pads with four teeth were fabricated to determine the number of turns and shapes for the transmitter and receiver based on the actual L_P, L_S, M, and R_ac. Figure 9 shows the actual wireless pads that were produced.

In each structure, the self-inductance L_P and L_S at 10 turns, 20 turns, 30 turns, and 40 turns are presented in

Table 3. Self-inductance according to number of turns and structure.

	No. Turns	10 Turns	20 Turns	30 Turns	40 Turns
Structure
Dipole		17.56 μH	65.64 μH	146.82 μH	280.93 μH
Multiple E		24.63 μH	89.41 μH	204.8 μH	352.52 μH

. As with the FEM simulations, it can be observed that the multiple E structure has a higher inductance compared to the dipole structure. This indicates that it can achieve a higher mutual inductance, allowing for a reduction in the primary coil current. However, due to the reduction in the window area caused by the teeth, the winding must be performed in a double layer, which may increase the proximity effect between the windings. The resistance of the coils based on different shapes and turns is presented in

Table 4. AC resistance according to number of turns and structure.

	No. Turns	10 Turns	20 Turns	30 Turns	40 Turns
Structure
Dipole		0.0264 Ω	0.1029 Ω	0.2301 Ω	0.2919 Ω
Multiple E		0.0321 Ω	0.1187 Ω	0.2613 Ω	0.4501 Ω

. An LCR meter from HIOKI (IM3536) was used for measurements at a frequency of 100 kHz. The dipole structure has a wider winding window area compared to the multiple E structure, allowing it to maintain a greater distance between windings, resulting in relatively lower AC resistance.

Next, the coupling coefficients were measured by varying the structures of the transmitter and receiver. As shown in

Table 5. Coupling coefficient according to pad structure combination.

	Secondary	Dipole	Multiple E
Primary
Dipole		0.037	0.041
Multiple E		0.041	0.046

, the coupling coefficients were measured for a total of four combinations. The dipole–dipole combination has a coupling coefficient of 0.037, indicating a very weak coupling. The dipole-multiple E combination shows an increased coupling coefficient of 0.041. Finally, the multiple E-multiple E combination has a coupling coefficient of 0.046, which represents a 24.32% increase compared to the dipole–dipole structure.

According to Table 3, Table 4 and Table 5, the multiple E structure exhibits higher self-inductance, coupling coefficients, and AC resistance compared to the dipole structure. Thus, according to Equation (5), while the high mutual inductance can reduce I_P, the relatively large AC resistance may lead to increased copper losses in the coil. Therefore, it is necessary to analyze the size of the copper losses based on the combinations, number of turns, and the structure of the receiver rectifier in this paper.

The electrical characteristics according to the number of turns in each combination are examined.

Table 6. Key parameters of dipole–dipole structure according to rectifier structure.

Symbol	Value
TPri	10				20
TSec	10	20	30	40	10	20	30	40
M (μH)	0.86	1.63	2.47	3.24	1.65	3.14	4.75	6.24
IP.Full (A)	167.34	87.83	-	-	86.92	45.62	30.14	22.97
IP.Half (A)	83.67	43.915	-	-	43.46	22.81	15.07	11.485
IS.Full (A)	5.55
IS.Half (A)	11.1
Symbol	Value
TPri	30				40
TSec	10	20	30	40	10	20	30	40
M (μH)	2.46	4.69	7.1	9.32	3.41	6.5	9.83	12.9
IP.Full (A)	58.18	30.54	20.18	15.38	42.01	22.05	14.57	11.11
IP.Half (A)	29.09	15.27	10.09	7.69	21.005	11.025	7.285	5.555
IS.Full (A)	5.55
IS.Half (A)	11.1

presents the parameters based on P_O = 500 W and U_DC = 100 V for the dipole–dipole structure with varying turns. Here, T_Pri denotes the number of turns in the transmitter coil, and T_Sec refers to the number of turns in the receiver coil. Additionally, IP.FBR indicates the transmitter coil current when using a full-bridge rectifier, while I_P.VDR represents the current size when using a voltage doubler rectifier. It can be observed that as the number of turns increases, the mutual inductance grows, resulting in a decrease in the primary coil current, which is influenced by both the primary and secondary coil turns. The current in the receiver coil is determined by the characteristics of the LCC-S topology and remains constant regardless of the number of turns, being 5.55 A for the full-bridge rectifier (I_S.FBR) and 11.1 A for the voltage doubler rectifier (I_S.VDR). If the mutual inductance is not sufficiently high, a very large I_P can be observed. When using the voltage doubler rectifier, I_P is reduced to half compared to the full-bridge rectifier, while I_S doubles. If the transmitter coil has 10 turns and the receiver coil has 40 turns, the designed C_F value based on Equation (8) becomes negative, making practical application impossible, as indicated by the dash. When using the receiver as VDR can be effectively reduced, suggesting that with a smaller number of turns, copper losses can be minimized. However, as the number of turns in the transmitter and receiver increases to ensure sufficient mutual inductance, I_S may become larger than I_P, indicating that copper losses in the transmitter become a significant proportion.

If the number of turns in the transmitter and receiver coils is insufficient, the resistance will increase, but it is evident that high efficiency can only be achieved when a sufficient number of turns are secured.

Table 7. Key parameters of dipole-multiple E structure according to rectifier structure.

Symbol	Value
TPri	10				20
TSec	10	20	30	40	10	20	30	40
M (μH)	0.86	1.65	2.47	3.41	1.63	3.14	4.69	6.5
IP.Full (A)	167.34	86.92	58.18	-	87.83	45.62	30.54	22.05
IP.Half (A)	83.67	43.46	29.09	-	43.915	22.81	15.27	11.025
IS.Full (A)	5.55
IS.Half (A)	11.1
Symbol	Value
TPri	30				40
TSec	10	20	30	40	10	20	30	40
M (μH)	2.47	4.75	7.1	9.83	3.24	6.25	9.32	12.9
IP.Full (A)	58.03	30.14	20.18	14.57	44.23	22.97	15.38	11.11
IP.Half (A)	29.015	15.07	10.09	7.285	22.115	11.485	7.69	5.555
IS.Full (A)	5.55
IS.Half (A)	11.1

shows the parameters for the dipole-multiple E structure based on the number of turns. The coupling coefficient increases compared to the dipole–dipole structure, indicating a larger M. Consequently, I_P is reduced compared to Table 6. However, due to the proximity effect arising from winding ferrite teeth in a confined space, the resistance value increases. Therefore, an analysis of combinations based on structure and the number of turns is necessary.

Table 8. Key parameters of multiple E-dipole structure according to rectifier structure.

Symbol	Value
TPri	10				20
TSec	10	20	30	40	10	20	30	40
M (μH)	1.13	2.16	3.27	4.29	2.16	4.11	6.22	8.17
IP.Full (A)	126.47	66.38	-	-	66.38	34.84	23.02	17.55
IP.Half (A)	63.235	33.19	-	-	33.19	17.42	11.51	8.775
IS.Full (A)	5.55
IS.Half (A)	11.1
Symbol	Value
TPri	30				40
TSec	10	20	30	40	10	20	30	40
M (μH)	3.27	6.22	9.42	12.36	4.19	8.11	12.39	16.21
IP.Full (A)	43.86	23.02	15.21	11.59	34.19	17.68	11.56	8.83
IP.Half (A)	21.93	11.51	7.605	5.795	17.095	8.84	5.78	4.415
IS.Full (A)	5.55
IS.Half (A)	11.1

presents the parameters for the multiple E-dipole structure based on the number of turns. It exhibits similar characteristics to Table 7, but due to the high I_P, it is advantageous to have a smaller winding resistance when having the same M. Thus, it can be concluded that this structure is less favorable compared to the combinations in Table 7.

Table 9. Key parameters of multiple E-multiple E structure according to rectifier structure.

Symbol	Value
TPri	10				20
TSec	10	20	30	40	10	20	30	40
M (μH)	1.13	2.16	3.27	4.29	2.16	4.11	6.22	8.17
IP.Full (A)	126.47	66.38	-	-	66.38	34.84	23.02	17.55
IP.Half (A)	63.235	33.19	-	-	33.19	17.42	11.51	8.775
IS.Full (A)	5.55
IS.Half (A)	11.1
Symbol	Value
TPri	30				40
TSec	10	20	30	40	10	20	30	40
M (μH)	3.27	6.22	9.42	12.36	4.19	8.11	12.39	16.21
IP.Full (A)	43.86	23.02	15.21	11.59	34.19	17.68	11.56	8.83
IP.Half (A)	21.93	11.51	7.605	5.795	17.095	8.84	5.78	4.415
IS.Full (A)	5.55
IS.Half (A)	11.1

shows the parameters for the multiple E-multiple E structure based on the number of turns. It has the highest mutual inductance among all cases, resulting in the smallest I_P. However, since the resistance of both the transmitter and receiver coils is the highest, the impact of this should be analyzed. Additionally, since coils generally account for the largest losses in IPT systems, it is necessary to analyze the conduction losses occurring in the coils.

Figure 10 presents the calculated copper losses of the transmitter and receiver coil based on a total of 32 combinations. Figure 10a shows the losses when using a full-bridge rectifier, while Figure 10b displays the losses when using a voltage doubler rectifier. When a full-bridge rectifier is used, the copper losses are significant at lower turns due to the high transmitter coil current, but it can be observed that the losses decrease as the number of turns increases with the reduction of I_P. Overall, the multiple E-multiple E combination exhibits the smallest losses, with the case of 40 turns for both the transmitter and receiver having the lowest copper losses. However, the combinations of T: 10/R: 40, T: 20/R: 40, and T: 30/R: 40 in the multiple E-multiple E structure also show similar loss levels. With the full-bridge rectifier, the multiple E structure is advantageous in terms of copper losses due to its ability to secure a higher self-inductance owing to the relatively high I_P current.

As shown in Figure 10b, when using the voltage doubler rectifier, I_P decreases to half compared to the full-bridge rectifier structure, resulting in an overall reduction in losses. As mentioned in Table 6, Table 7, Table 8 and Table 9, cases where the C_F value results in a negative value are not feasible for actual fabrication. Therefore, these cases are not included in the figure. Due to the reduced I_P current, the influence of the increase in self-inductance leading to a decrease in current is less significant than in the full-bridge rectifier structure, resulting in reduced losses for all combinations.

Based on these results, three combinations were selected. The first is the multiple E-multiple E structure with T: 40/R: 40 for the full-bridge rectifier. The second is the multiple E-multiple E structure with T: 20/R: 40 using the voltage doubler rectifier. Although other turn counts yield similar loss magnitudes within the same structure, this combination was chosen due to its lower number of turns, which is beneficial in terms of overall weight and cost. Lastly, T: 20/R: 20 was selected for the dipole-multiple E structure using the voltage doubler rectifier to allow for comparison of this configuration with the others based on the calculated results.

The design procedures performed in Section 3 have been summarized in Figure 11. Selecting the optimal pad combination should prioritize minimizing conduction loss in the coil. However, when considering system cost and weight, a combination with lower turns may be selected, even if it results in a slight increase in conduction loss.

4. Experimental Validation of the Proposed System

In this paper, a system capable of 500 W and 100 V wireless power transfer over a distance of 250 mm was fabricated for the three cases analyzed in Section 3. Based on the parameters of the wireless pad, L_IN, C_P, C_F, and C_S were designed according to Equation (8). The designed resonant parameters are summarized in

Table 10. Value of compensation network parameters according to pad structures.

Symbol	Case I	Case II	Case III
LIN (μH)	59.58	44.23	22.35
CP (nF)	42.51	57.27	113.34
CF (nF)	8.73	57.61	59.38
CS (nF)	7.20	12.35	28.40

. Here, Case I refers to the combination of multiple E-multiple E with T: 40/R: 40, Case II refers to T: 20/R: 30, and Case III refers to the combination of dipole-multiple E with T: 20/R: 20. The switches used in the inverter are CREE’s C3M0030090K, and the diodes used in the rectifier are Infineon’s IDW30G65C5. And components of magnetic cores in the resonant inductor, wireless pad, and resonant capacitor are listed in

Table 11. System key components list.

Component	Name	Manufacturer
Inverter MOSFET	C3M0030090K	CREE, Durham, NC, USA
Rectifier diode	IDW30G65C5	Infineon, Neubiberg, Germany
LIN magnetic core	PQ5050 (PM16)	TODA ISU, Gangwon, Republic of Korea
Pad magnetic core	90 × 100 × 10 mm (PM11)	TODA ISU, Gangwon, Republic of Korea
Teeth magnetic core	100 × 37.5 × 10 mm (PM11)	TODA ISU, Gangwon, Republic of Korea
Resonant capacitor	C3640C153KGGACAUTO, C2225C392JGGACTU, C1812C222KDGACTU	KEMET, Fort Lauderdale, FL, USA

.

Figure 12 shows the fabricated transmitter and receiver pads for the three cases, with the addition of aluminum shielding. Figure 13 presents the key waveforms for each of the three cases when the system operates at 500 W and 100 V output. As shown in Figure 13, zero voltage switching (ZVS) operation is confirmed in all cases. In Figure 13a, with the use of a full-bridge rectifier, the peak value of U_ab is confirmed to be 100 V, whereas in Figure 13b,c, U_ab has a peak value of 50 V.

Figure 14 illustrates the voltage, current, and power levels of each component measured using a power analyzer. Case I, which has the highest mutual inductance, exhibits the smallest I_P value. It can be observed that Cases II and III, which use a voltage doubler rectifier, have I_S values that are twice as large as those of Case I, which uses a full-bridge rectifier. When the output rectifier is the same, except for I_P, the current and voltage levels of each component are almost identical, indicating that the system efficiency is primarily affected by copper losses. Therefore, as analyzed in Section 3, Case II, which has the smallest copper losses, achieves the highest efficiency of 80.948%, while Case I has an efficiency of 76.717% and Case III has an efficiency of 71.145%.

Figure 15 shows the efficiency variation in each case based on the load level. It can be seen that case II has the highest efficiency across all load ranges. Due to the characteristics of the LCC-S structure, as the load changes, the receiver current I_S varies according to Equation (2), but the transmitter current I_P remains constant by Equation (1). Therefore, in case III, where mutual inductance M is smaller, the efficiency significantly decreases when the load is low. The experimental results indicate that the combination derived from the proposed design method achieves the highest efficiency. The loss distribution analysis for Case I, II, and III is performed based on the measured efficiency at 500 W, as shown in Figure 16. In Case III, the current applied to the I_P is larger than in other cases, resulting in the highest loss in the IPT pad. Although the rectifier diode with VDR shows an increased current compared to FBR, the total number of components used is halved, resulting in negligible differences. The IPT pad losses in Case I and Case II account for similar proportions; however, the actual value in Case I is higher due to the increased coil resistance. Since the losses generated in the coil constitute the largest proportion in all cases, the process of calculating the conduction loss of the coil with the lowest value, as described in Section 3, is considered appropriate.

Experiments were conducted to verify the robustness of the system under misalignment and switching frequency variations.

Dynamic characteristic experiments for load variations are added for Case II under x-axis misalignments of 10 mm, 30 mm, and 50 mm, as shown in Figure 17a–c, respectively. For a 10 mm misalignment, a power transmission reduction of 17 W is observed under maximum load conditions. In other conditions, the power transmission reduction can be negligible. At a 30 mm misalignment, 428 W of power was transmitted at a load condition of 19.25 Ω, which corresponds to 500 W under alignment. At a 50 mm misalignment, 358 W of power is delivered to the load. In this study, the IPT converter is followed by a DC-DC converter. Therefore, in practice, the reduced power transmission can be compensated through DC-DC converter control. Additionally, the system efficiency of Case II under different misalignments was included, as shown in Figure 18. The overall efficiency decrease is not significant up to a 30 mm misalignment. Notably, under a 10 mm misalignment, the efficiency at 25 Ω and 35 Ω load conditions are similar to that of the aligned condition.

The output voltage regulation performance by load fluctuations for Case II were progressed, as shown in Figure 19. It was observed that the output voltage remained stable with only a slight voltage drop due to the coil resistance. In this case, no additional output control was applied, and a constant switching frequency of 100 kHz was used.

Experiments were also conducted to validate output variations at the same load resistance under reduced switching frequencies of 98, 96, 94, and 92 kHz, as shown in Figure 20. It was observed that output power significantly decreased with frequency reduction, and at a switching frequency of 96 kHz, the output power dropped below 100 W under the same load resistance. This demonstrates that, for the proposed system, the switching frequency should follow the resonant frequency to achieve maximum power transfer.

5. Conclusions

This paper presented the design, implementation, and experimental validation of a 500 W inductive power transfer (IPT) system for television applications, capable of achieving wireless power transfer over a 250 mm distance. The proposed system focused on optimizing the wireless pad design using a four-teeth magnetic structure to enhance coupling coefficients and reduce copper losses. Through a detailed analysis of pad structures and coil configurations, the optimal combination was identified, balancing efficiency, cost, and weight considerations. Experimental validation demonstrated that the multiple E–multiple E structure with a voltage doubler rectifier achieved the highest efficiency of 80.948%, highlighting the effectiveness of the proposed design approach. The system also exhibited robustness under misalignment conditions, maintaining high efficiency up to a 30 mm x-axis displacement. Additionally, the importance of maintaining resonance conditions was emphasized, as switching frequency variations significantly affected power transfer performance. Overall, the proposed IPT system addresses critical challenges in long-distance wireless power transfer, providing a reliable and efficient solution for high-power applications. The findings contribute to advancing wireless power transfer technology, particularly for home appliances, and lay the groundwork for future research on further improving system performance and integration.

6. Discussion

The experimental and simulation results confirm the effectiveness of the proposed 500 W IPT system for a 250 mm transmission distance. The results demonstrate several key findings:

• Coupling coefficient and magnetic path design: The use of a four-teeth magnetic structure significantly enhances the coupling coefficient compared to conventional designs, such as circular or dipole types. This improvement directly translates to reduced primary coil current (I_P) and overall system losses, as evident in the analysis presented in Section 3. The inclusion of teeth allows for better flux alignment, which is critical for long-distance wireless power transfer.
• System efficiency and loss analysis: The multiple E–multiple E structure with a voltage doubler rectifier achieved the highest efficiency (80.948%), attributed to its optimal balance of mutual inductance and copper losses. However, Case III, which utilizes a dipole–multiple E structure, exhibited reduced efficiency due to lower mutual inductance and higher resistance, reinforcing the importance of proper pad design and coil configuration.
• Robustness to misalignment: The system demonstrated robustness under x-axis misalignments of up to 30 mm, with efficiency reductions being minimal. Beyond this threshold (e.g., 50 mm misalignment), the efficiency and power transmission significantly decreased, highlighting the need for precise installation to maintain performance.
• Frequency sensitivity: The results show that the system’s output power is highly sensitive to switching frequency variations. When the switching frequency deviated from 100 kHz, the power transfer efficiency dropped sharply, particularly at 96 kHz, where output power fell below 100 W. This underscores the importance of maintaining resonance conditions for optimal operation.
• Load variation performance: Under load fluctuations, the system maintained stable output voltage with minimal variation, except for minor drops caused by coil resistance. This stability, achieved without additional output control, demonstrates the system’s potential for practical applications in TV appliances.

Overall, the proposed system addresses key challenges in long-distance wireless power transfer, including high efficiency, misalignment robustness, and stable operation under varying conditions. The experimental validation supports the practicality of the design, making it a promising solution for high-power, long-distance wireless power transfer in home appliances.