Online Failure Diagnostic in Full-Bridge Module for Optimum Setup of an IGBT-Based Multilevel Inverter

Abstract

An online failure diagnostic test is essential to ensure the robustness and reliability of high-powered systems. Furthermore, the overall design must comprise diagnostic strategies to detect in-service and high-powered module defects. This paper describes the critical failure mechanisms––cross-conduction, inductive avalanche, second turn-on, VS-undershoot, inrush current, and thermal runaway––that directly affect insulated gate bipolar transistor (IGBT) operation. The constructed inverter contains 18 transformer-based taps (six per phase); however, this work studied a single tap (IGBT-based full-bridge module) to understand the reasons for failure and the routes to mitigate them. Moreover, a cost-effective solution using the IR2127STRPBF driver circuit was implemented to reduce the probability of thermal runaway in case of overcurrent, short-circuit, or avalanche events. For this reason, the electrical current state was adjusted using an FPGA digital resource to perform dynamic PWM control signals. The obtained correlation waveforms are valuable for verifying diagnostic data at the design stage to emphasize the significance of evading premature failure events. The comprehensive study on failure diagnosis enabled successful design strategies to construct a robust 45 kVA three-phase multilevel inverter for a 22 kW eolic-photovoltaic generation plant.

1. Introduction

Multilevel inverters (MLIs) have become more popular than conventional DC/AC inverters for some application fields because they operate with many voltage steps that improve the output waveform. They are categorized into non-isolated and isolated multilevel inverters, which are widely used in renewable energy systems and industrial applications, among others. Isolated MLIs exploit low-frequency transformers to improve robustness and reliability and are subcategorized into DC-side and AC-side. The MLIs feature several advantages over conventional topologies. MLIs operate at both high and low frequencies, reducing the device-switching rate; feature low-output harmonic distortion; achieve a high-power capability by dividing the total power into several switching devices; and extend the switching device’s lifetime and reliability due to low switching rates. Low-frequency MLIs reduce switching losses considerably, thereby enhancing overall efficiency. An extensive comparison among multilevel topologies (state of the art) elaborates on [1].

In this work, the low-frequency transformer plays a critical role at the design stage. Its electrical properties, such as high inductance, high inrush current, reduced di/dt, and high efficiency at heavy loads, determine several design techniques to prevent failure [2,3,4]. In addition, the full-bridge module sets up two arms that contain two isolated gate bipolar transistor (IGBT) devices. In any situation, one IGBT within an arm is solely on-state. However, when both IGBTs are on-state in a failure scenario, the current through the IGBT devices abruptly increases. This current is, in most cases, high enough to instantly damage the switching devices. In addition, the enormous di/dt may damage the IGBT and driver circuit, as seen in Figure 1. This failure mechanism is called a “type I” short-circuit event. However, if the event comes from the transformer output, the current flows through the transformer primary winding, reducing the di/dt considerably in the IGBT devices. A reduced di/dt enables a higher time window within the IGBT that must be turned off before firing. This failure scenario, which originated externally, is called a “type II” short-circuit event, which has a higher occurrence probability than type I. Figure 1 illustrates these possible short-circuit scenarios. Here, the circuit design must consider both failure events to prevent IGBT damage by turning it off immediately.

This work focuses on online failure diagnostics in AC-side isolated inverters, especially the cascade transformer multilevel inverter (CTMI). The inverter is part of an overall project of sustainable energy generation and distribution within a region where electric power lines are unavailable. The energy generation plant consists of a 12 kW solar panel generator, a 10 kW wind turbine, a 240 V-435 Ah battery bank, an isolated 45 kVA three-phase multilevel inverter, and a power distribution box.

The full-bridge module design is critical for the optimum performance of an energy generation plant. The failure mechanisms in the IGBT-based full-bridge module, such as cross-conduction, inductive load avalanche, second turn-on, VS-undershoot, and thermal runaway, will be described in Section 2. Measurements of the online failure mechanisms in a single-power stage (one full-bridge module) will be covered in Section 3. Section 4 discusses the transformer’s influence on current conduction and IGBT operating temperature when a driver circuit is used for online diagnostic as a function of the electrical current state. Using pulse width modulation (PWM) in a field programming gate array (FPGA) will also be detailed in this section. Finally, the conclusion is presented in Section 5.

2. Failure Mechanisms and Operation of a Full-Bridge Module

CTMIs are extensively used in large solar power plants because they feature low harmonic distortion, galvanic isolation, and low-frequency switching operation [5]. Moreover, CTMIs can improve reliability, robustness, and efficiency, despite the use of low-frequency transformers with high inductance [6,7,8,9]. More importantly, using a DC source solely has been cited as a CTMI advantage. The main disadvantage, however, is that it requires large transformers, which implies increased weight and inverter size.

As shown in Figure 1, the modular CTMI topology involves two types of short circuits. Short-circuit type I occurs when the current travels across a single arm of the full-bridge, whereas type II occurs when the current passes through the transformer’s primary winding. Several techniques to prevent IGBT failures have been published, e.g., [10]. However, many of these techniques are difficult to apply; moreover, they are not fully developed. The key critical failure mechanisms found on each full-bridge module incorporated at the CTMI are discussed below:

• The cross-conduction mechanism emerges during the switching intervals while two IGBTs of the same arm are active. It means an IGBT turns on while the opposite IGBT has not yet turned off. Higher IGBT turn-off times and the driver propagation delays originate from this phenomenon. The cross-conduction is directly related to the switching losses; the higher the cross-conduction, the higher the loss. Furthermore, excessive cross-conduction produces high collector current spikes during transitions that gradually degrade the switching device. IGBT degradation leads to failures such as bonding-wire cracking and detachment, die stresses, and in some cases thermal breakdown.
• In high-inductance switching applications, managing the inductor-stored energy properly is a challenge nowadays. This stored energy becomes a high voltage spike that appears during the IGBT turn-off transition, causing an inductive load avalanche failure. The avalanche mechanism deals with the energy in a different way than a short-circuit does. In a short-circuit event, the IGBT is active, whereas in the avalanche event, the IGBT is gate-emitter voltage (VGE) unbiased.
• The second turn-on is a failure mechanism originated by the IGBT’s Miller capacitance combined with a high dv/dt at the collector node and leads to a type I short-circuit failure. This mechanism occurs when the high voltage at the collector node induces a voltage spike in the gate node during the IGBT turn-off transition. If the voltage spike reaches the IGBT plateau level, the device will turn on again. In addition, the induced voltage in the gate node may increase enough to reach the plateau level, turning on the IGBT, see Figure 2.
• The VS-undershoot failure mechanism is an under-voltage spike that appears at the driver’s VS terminal, which is used for IGBT switching. When a high di/dt transition occurs, it is generated by the stored energy in the parasitic inductances of the collector and emitter IGBT terminals of the PCB tracks (Lstray). Therefore, VS-undershoot directly affects the high-side IGBT driver, as seen in Figure 3.
• Thermal runaway is a failure mechanism that occurs when a rising temperature causes a further increase in temperature in each IGBT in the full-bridge module. Therefore, uncontrolled temperature-positive feedback leads the IGBT to a destructive scenario.

Figure 4 presents a constructed 45 kVA AC-side-isolated multilevel inverter block diagram. It shows a modular CTMI of 18 transformer-isolated power modules, FPGA-based control signal processing, microcontroller-based analog signal processing, human machine interface (HMI), analog front-end, DC main switch and protections, control-signal isolation module, isolated low-voltage power supplies, voltage and current sensors, DC-link, and control signals distribution. A single power stage called “single tap” is equivalent to the full-bridge module and comprises four IGBTs and a 2.5 kVA low-frequency transformer (see Figure 1). Here, failure mechanisms come from this module.

Previous literature has revealed several design techniques to detect short-circuit events. For example, conducting power semiconductor transistors, such as the metal oxide semiconductor field effect transistor (MOSFET), turn off quickly by adding a current-sensing circuit into the drain terminal [11]. This technology has advantages over the conventional shunt method because it does not directly affect the load voltage. Nevertheless, these special switching devices are not widely available commercially.

A short-circuit detection method based on current shunts built into an IGBT power module that effectively reduces the IGBT turn-off time upon a short-circuit event has been introduced [12]. However, that method uses shunt resistors that require additional circuit stages, such as an RC compensation circuit and a filtered amplifier. Other methods use di/dt and dv/dt detection to infer a short-circuit event [13,14]. Nonetheless, these methods require complex hardware. Finally, another method analyses the IGBT gate voltage to detect hard switching faults and faults under load [15]; the IGBT driver hardware is significantly more complex.

This authors of this manuscript prefer conventional IGBT desaturation to previous methods for online diagnostic short-circuit events for its robustness, well-known trade-offs, and commercial availability of the IGBT driver. However, the IGBT desaturation technique involves a significant IGBT turn-off time on a short-circuit event. This is called the “blanking time (tb)”, which depends on a passive integrator RC constant and a propagation delay of each implemented IGBT driver circuit. A large tb combined with a high desaturation current could lead to an IGBT thermal breakdown. References [16,17,18,19] explain how to improve the desaturation technique by reducing tb even if the IGBT operates with high currents. Therefore, it determines the short-circuit time that the IGBT must withstand. Figure 5 depicts the main components of the fabricated full-bridge module (single tap). The critical design specifications are listed in

Table 1. Design specifications of the implemented full-bridge module (single tap).

Specifications	Values
Input voltage (DC link)	230–280 V (240 V nominal)
Output voltage (square wave)	230–280 V peak (variable duty cycle)
IGBT switching frequency	60 Hz (low switching loss)
Output THD	<8%
Shor circuit current limit	300 A
Overload (15 min)	150%
Power PCB board size (L × W)	180 × 70 mm

.

3. Online Diagnostic of the Failure Mechanisms

The circuit design proposed here for the online diagnostic of failure mechanisms due to short-circuit events improves on the one proposed in [20]. Figure 6 depicts the enhanced design of the high-side driver solely. The principal changes comprise the driver substitution by the IR2127STRPBF integrated circuit, the gate voltage clamp, the VS-undershoot reduction resistor (R4), and the use of a short-circuit tolerant IGBT.

According to [21], R2 is typically chosen to be 10 k for a bootstrap voltage V_B = 12 V, 22 k for V_B = 15 V and 33 k for V_B = 18 V. The voltage V_X is typically 9.2. All the component values of the proposed circuit are listed in

Table 2. Bill of material for the applied short-circuit current diagnostic topology.

Device Symbol	MFG Part	Description
R1	RC0603FR-075K6L	Thick Film Resistor 5.6 kΩ-1%
R2	CRCW060310K0JNEAC	Thick Film Resistor 0.1 W 10 kΩ 5%
R3	CRCW0603200RFKEAC	Thick Film Resistor 0.1 W 200 Ω-1%
R4, RG	CR1206-FX-10R0ELF	Thick Film Resistors 10 Ω-1%
R5	CR0603-JW-392ELF	Thick Film Resistor 3.9 kΩ-5%
C1	TAJC476M016RNJ	Tantalum capacitor SMT, 47 µF, 16 V
C2	CC0603KRX7R9BB103	Multilayer Ceramic Capacitor MLCC-SMD/SMT 10 nF 50 V X7R 10%
C3, C4	CC0603ZRY5V9BB104	Multilayer Ceramic Capacitors MLCC-SMD/SMT 0.1 µF 50 V 20–80%
D1, D2, D4	ES1J	Super-Fast Diode SMA 600 V 1A
D3, D6	MM4148-T	Diode-General Purpose, Power, Switching RECT-SWIT 0075V 00.15 A
D5	ZMM5245B	Zener Diode 0.5 W 15 V 5%
DC-LINK	B43504A9567M000	CAP ALUM 560 µF 20% 400 V SNAP
DECOUPLING	ECW-F4205JL	CAP FILM 2 µF 5% 400VDC AXIAL
Q1	NGTB50N120FL2WG	1200 V/50 A FAST IGBT FSII

. These components are jointly optimized to achieve stable performance across the operation conditions. This optimization was performed by several trade-offs of the relevant component values as follows:

• R1, R2, R3, and C2 values established the driver shutdown delay to prevent an unstable failure condition during the IGBT turn-on time. Therefore, the minimum shutdown delay was the IGBT turn-on time, and the maximum was the IGBT short-circuit withstanding time minus the driver shutdown propagation delay. The optimum shutdown delay should be slightly higher than the IGBT turn-on time.
• The RG value predominantly sets the gate impedance, which should be as low as possible to reduce the probability of the second turn-on phenomenon. However, the RG also creates a delay path to increase the IGBT turn-on time, preventing cross-conduction failure. Moreover, it slightly increases dv/dt, reducing the harmful effects of the Miller capacitance. The optimum RG value is the one that equals the IGBT turn-on and turn-off times.
• The boost capacitor C1 stores the energy to keep the high-side IGBT active when needed. Its value is essential, especially in low-frequency applications. C1 also plays an essential role in setting the driver’s output impedance during the high logical state. In Section 3.3, detailed information on the C1 optimum value calculation is presented.

The circuit in Figure 6 sets up R1, R2, R3, C2, D1, and the IR2127STRPBF driver to accomplish the online diagnostic on the current signal. The operation principle is based on the statement that a saturated IGBT (Q1) increases its collector-emitter voltage (V_CE) as the collector current increases. The higher the I_C, the higher the V_CE. The IGBT driver uses a current sense (CS) terminal to detect an I_C increase by reading the V_CE voltage.

When Q1 is inactive, I_C is close to zero, and V_CE equals the DC-link power supply voltage (V_BUS). At this stage, D1 isolates the collector voltage of the voltage divider built with R1 and R3. The HO terminal remains low; thus, CS voltage goes below the threshold voltage V_CSth [22], determining a regular operation. If the HO terminal asserts that V_CE remains equal to V_BUS until the turn-on time elapses, the CS voltage gradually increases due to the integrator action of R1, R2, R3, and C2. CS voltage must not reach V_CSth before V_CE goes down to prevent an erratic short-circuit event. However, when V_CE finally goes down, D1 becomes active and establishes a V_CS–V_S voltage below V_CSth.

If a short-circuit event occurs while Q1 is active, V_CE increases to a value determined by the desaturation current and the V_GE voltage [23]; therefore, D1 is reverse-biased and isolates the CS voltage divider of the collector voltage. As the HO terminal stills asserted, CS starts rising again until it reaches V_CSth. At this point, the CS signal is propagated into the driver until the HO signal shutdowns the IGBT and the FAULT terminal is driven low, indicating a short-circuit failure event.

After building the improved IR2127STRPBF driver, it was implemented into the power board as shown in Figure 5, and an IGBT-based multilevel inverter module was tested.

Figure 7 shows the measurement setup, where all measurements were taken in the laboratory under fault-tolerant scenarios. Three 700 W resistive loads, a Xilinx Zynq-7000 FPGA board, an isolation interface between the FPGA and the power board, and a 2.5 kVA low-frequency transformer were used as the testbench. Furthermore, a 10 kVA three-phase multi-tap isolation transformer for security purposes was also used.

Failure-event measurement on the IGBT-based full-bridge inverter (single tap) was performed and evaluated by correlation plots of the electrical parameters. These measurements required a digital-storage oscilloscope type Siglent SDS-1102CML (Siglent Technologies, Shenzhen, China) with low-capacitance probes (model P6120: Tektronics Inc., Beaverton, OR, USA). Current values were measured using an AC clamp (model i400s: Fluke Corporation., WA, USA), Magneto-resistor current sensor (model DS10.2s: Tell-i Technologies Inc., Charlotte, NC, USA), and a 1000 A bidirectional current sensor (model SSA-1000: Riedon Resistors., Alhambra, CA, USA) and measured when connected to a Siglent SDS-1102CML oscilloscope (see Figure 7).

A technical description related to each failure mechanism evaluated in the “power board” is given below.

3.1. Cross-Conduction

Since the low-side IGBTs of the full-bridge module also operate as clamping devices for the high-inductive load of the transformer’s primary winding, it is necessary to keep a certain level of cross-conduction to prevent an unclamped high-inductive scenario. A good trade-off equals the IGBT turn-on and turn-off times. Passive delay lines equal the switching times to achieve a good balance between switching losses and unclamped high inductive spikes (UIS). For example, RG, D3, R5, C4, and D6 from Table 2 set up two delay lines to control cross-conduction. Figure 8 shows the experimental results from measuring the low-side and high-side switching times. After the passive-delay lines design, turn-on and turn-off times are almost equal.

Figure 9a depicts the high-side V_CE and I_C, waveforms. Notice that the IGBT lies within the safe operating area (SOA) where cross-conduction does not visually influence the collector current, and V_CE does not present high-voltage spikes due to UIS. Figure 9b shows the low-side V_CE and I_C waveforms. The low-side turn-on transition lies within the clamping stage of the transformer’s primary winding inductance. However, a slightly negative clamping current spike was observed.

3.2. Inductive Avalanche

Under this condition, the current distribution across the surface of the IGBT die was uniform, maximizing the ability to dissipate the energy. In the avalanche event, the IGBT was non-conducting with no bias on the gate. Thus, when an avalanche occurred, the current travelled around the perimeter of the die, creating a non-uniform current distribution across its surface, so the maximum current for failure was much less in the avalanche event than in the short-circuit event [24,25].

Because of high transformer inductance, several precautions had be taken: (1) dv/dt was limited by increasing the R_G value. This change did not affect conduction losses as V_GE did; (2) A certain level of cross-conduction was permitted; however, the optimum cross-conduction time was zero; (3) switching coding was used to let the inductive load clamp through the low-side IGBTs; (4) the printed circuit board (PCB) parasitic inductances (L_stray) were reduced (5) The equivalent series resistance (ESR) and equivalent series inductance (ESL) of the DC-link capacitors were reduced. As seen previously, dv/dt was reduced. Notice that the low switching frequency operation did not influence efficiency because of switching loss. According to [26], dv/dt was limited by physical parameters, such as threshold gate voltage (V_TH), total gate resistance (RG), and gate-emitter capacitance (C_GC). Thus, the ESR was reduced by paralleling low ESR capacitors as closely as possible to the IGBTs.

Measurements of V_CE, I_C, and V_GE were taken during the IGBTs turn-off transition of high- and low-side IGBTs accordingly. Figure 10a shows the V_CE, I_C, and V_GE waveforms during the turn-off transition of the high-side IGBT. Notice that V_CE did not observe voltage spikes, and the IGBT tail current was disclosed. Figure 10b shows the V_CE, I_C, and V_GE waveforms during the turn-off transition of the low-side IGBT. Again, V_CE transition did not present spikes from the clamping process implemented by programming. Again, the low-side IGBT turn-off transition occurred during the clamping time, so no further I_C transition was observed.

3.3. Second Turn-On

Gate–node impedance and dv/dt strongly influenced the gate-emitter voltage; therefore, voltage induction was reduced by adding an external capacitor between the gate and emitter nodes [27]. Gate impedance was reduced by paralleling an ultra-fast-recovery diode to RG. As a result, RG was obtained according to [28] and equaled 10 Ω.

Figure 11 presents the influence of dv/dt during the turn-on transition. Its influence on V_GE was also seen during the turn-on transition, thus avoiding the risk of causing a second turn-on event due to induced spikes through the scope probes. After decreasing gate impedance, a small influence was observed, and V_GE was poorly affected by dv/dt.

3.4. VS-Undershoot

Since CTMIs use low-frequency transformers that limit di/dt, one would think that VS-undershoot could be neglected. However, during a short-circuit event or an overload condition, the L_stray current could be high enough to undershoot the VS terminal. Thus, the high-side driver could fail. Reference [29] suggests some general recommendations to minimize VS-undershoot: (1) use a thick, direct track between switches to avoid loops or deviations; (2) avoid interconnect links; (3) consider re-locating IGBTs to reduce track lengths; (4) connect VS and COM as close as possible to the IGBT pins; (5) locate the driver as close as possible to the IGBTs; (6) increase the bootstrap capacitance using low-ESR capacitors; (7) use a second low-ESR capacitor from VCC to COM; (8) add a resistor R4 from COM to the IGBT emitter; and (9) add a fast recovery diode D4 from COM to VS, as shown in Figure 6. Here, the VS-undershoot severity is measured as the difference between (V_B − V_S) − (V_S − COM) under overload or short-circuit conditions.

Figure 12 shows the VS-undershoot phenomenon under an overload condition. Isolated probes were required to accomplish this measurement. The condition met the following parameters: power = 3.52 kVA; transformer secondary winding current I_SEC = 176.3 A; overload rate = 150%; and room temperature = 31 °C, respectively. The VS-undershoot severity was neglected after following the previous recommendations. Furthermore, the VS-undershoot directly affected the high-side driver leading to a latch-up failure.

3.5. Thermal Runaway

A method to predict the IGBT junction temperature and negative impact of high temperature on the die solder joints is introduced in [30]. A high collector current mainly causes thermal runaway during turn-off transition [31]; therefore, the following design considerations should be taken to prevent this condition: (1) Limit the short-circuit current by decreasing V_GE but not so much that it causes excessive power losses, increasing the device temperature; (2) Reduce the short-circuit “blanking time (t_b)” as much as possible; (3) Keep VB ripple voltage as low as possible by installing a bootstrap capacitor; (4) Maintain the IGBT case temperature as low as possible by using a robust thermal management system. Thermal management was not described in this paper; however, the results included IGBT temperature under several load conditions. Figure 13 presents three conditions when the IGBTs were mounted on a 180 × 130 × 33 mm aluminum heat sink without force convection. In addition, the average temperature of the four IGBT cases was reported.

4. Discussion

In low-frequency switching applications, a high transformer inrush current at the start may affect the DC-link voltage, leading to an input voltage failure, and the bootstrap capacitor in the driver circuit IR2127STRPBF plays a critical role in full-bridge module efficiency. Under this scenario, conductive losses were higher than switching losses because large power transformers featured nonlinear core magnetization, and a relatively low V_GE implied a higher V_CE, increasing IGBT temperature.

The modular CMIs contain several transformers, so the total inrush current drawn from the DC-link was the sum of each transformer’s particular inrush current. As a result, a huge current drawn from the DC-link led to failure due to a transient voltage drop. Indeed, a good selection of DC-link capacitors is central to preventing design drawbacks, and large bootstrap capacitors are required to prevent a considerable voltage drop in V_GE during conduction to keep ripple voltage V_B as low as possible [32].

Figure 14 shows a dynamic correlation between inrush current and voltage drop at the IGBT’s collector terminal when a 47 µF low-ESR bootstrap capacitor was used. For this purpose, a tantalum capacitor was connected in parallel with a ceramic capacitor. The PWM technique at the inverter startup considerably reduced the inrush current to prevent DC-link failure. Figure 14 also includes the transformer’s inrush current waveform without a PWM soft-start process.

The IGBT in the full-bridge module can withstand a collector current of 300 A during the blanking time without suffering failure. The maximum collector current was controlled, and the short-circuit current was limited mainly by V_GE. An excessive collector current during turn-off can lead the IGBT to either a latch-up mechanism failure or thermal runaway when a higher I_C and V_GE nominal value is 13 V. In any case, the IGBT would be destroyed.

Likewise, a reduced di/dt can be observed due to the high inductance of the transformer’s primary winding. Here, the collector current was restricted by V_GE and the CS voltage divider in the IR2127 driver. Thus, the designer had to ensure that the product of V_CE and I_C during desaturation lay within the safe operating area (SOA).

Figure 15a shows the fault terminal voltage (V_FL) when a high di/dt short-circuit failure mechanism occur, and that the total blanking time after the IGBT shutdown process began was around 2 µs. Figure 15b shows the short-circuit failure mechanism when the transformer secondary winding current (I_SEC) caused a short-circuit around 800 A, whereas the transformer primary winding inductance limited di/dt. In this case, the blanking time resulted in 470 µs for a V_GE = 13 V.

Additionally,

Table 3. Comparison of different methods for detecting and processing failures to improve IGBT performance.

Feature	Juan Carlos Iglesias-Rojas, et al.	Zhaoliang Meng, et al. [12]	Johannes Groeger, et al. [14]	Min Chen, et al. [17]	Shan Yin, et al. [18]
Method	Online monitoring + SC tolerant IGBT	Integrated shunt method	dv/dt and di/dt sensing	Self-adaptive blanking circuit	Improved VCE desaturation
Complexity	Low	Medium	High	High	Medium
Mature technology (reliability)	Yes	No	No	No	Yes
Blanking time	2 µs	380 ns	Not reported	500 ns	910 ns
Economic impact	Low Field Stop II IGBT	Medium Shunts Detection circuits	High High-speed ADCProcessor device	High Complex blanking circuit	Medium Special IC driver

resumes some critical features of this work compared with recent research for improving IGBT performance.

5. Conclusions

The measurements focused on online failure detection when an isolated full-bridge inverter module operated for more than 20 months under adverse conditions confirmed that this research project had a reliable and robust design. By using two passive delay lines inserted into the control signals path to prevent cross-conduction successful failure-event reduction was achieved because of a meticulous design trade-off between cross-conduction and proper transformer inductance energy deviation, which prevented an inductive load avalanche. Furthermore, a low IGBT gate impedance prevented a second turn-on occurrence because of the Miller capacitance and high dv/dt. Proper PCB design techniques lowered the VS-undershoot caused by parasitic inductances of the collector and emitter IGBT terminal paths. We also showed the importance of thermal management because a relatively low IGBT package temperature reduces the probability of thermal runaway in case of overcurrent, short-circuit, or avalanche events. Hence, online failure diagnostics solved by using cost-effective integrated circuits can ensure an efficient, reliable, and robust operation of high power systems under critical failure events.