Topology and Control Strategy of Multi-Port DC Power Electronic Transformer Based on Soft Switching

Abstract

Multi-port DC power electronic transformer (PET) is a core equipment for achieving transformation of different voltage levels and flexible interconnection of different DC buses in a DC distribution system. It is capable of bidirectional energy flow, flexible regulation of power flow, port fault isolation, and other functions. A new five-port DC transformer topology based on soft switching technology is proposed in this paper. In this topology, different DC voltage levels can be interconnected efficiently, such as 20 kV, 750 V, ±375 V, and 300 to 500 V adjustable. The control of each port is simple and flexible. The output voltage is stable, and they are independent of each other, which can improve the system reliability. The topology of the proposed multi-port DC transformer is introduced in detail. The working principle, control strategy, and parameter design method of the transformer are analyzed. Simulations and experimental results are provided to validate the theoretical analysis.

1. Introduction

With the intensification of energy crisis and environmental issues, global energy consumption is gradually shifting from traditional fossil fuels to distributed energy sources such as photovoltaic, wind power, and fuel cells [1,2]. The distributed generation system based on DC distribution network is an important component of the future new power system. Compared with traditional AC distribution networks, there are no reactive power and synchronization issues in the DC distribution systems, and it also has advantages such as low losses and less wire usage.

Nowadays, DC ports with different voltage levels are required in many applications, including hybrid/fuel cell vehicles, renewable energy, water electrolysis, etc. [3,4,5]. A multi-port DC transformer is an electrical device with multiple ports, which can directly connect distributed power generation systems, energy storage devices, and load. The device integrates multiple single-stage converters, achieves electrical isolation in topology, and can achieve multi-directional energy flow and multi-voltage level outputs in control [6,7,8,9].

A typical PET topology is a three-stage structure, including a rectifier circuit, isolated DC/DC circuit, and inverter circuit [10,11]. In [12,13,14], a three-stage topology based on cascaded H-bridge (CHB) has been proposed, which mainly includes an AC/DC rectifier with cascaded H-bridge modules, an output-parallel DC/DC converter, and a two-level inverter. The three-stage PET based on CHB has three ports, which include a middle-voltage AC (MVAC) port, low-voltage DC (LVDC) port, and LVAC port. The LVDC port can serve as an interface for distributed energy sources such as photovoltaic, energy storage, and DC loads. The medium-voltage port and low-voltage port of the PET is isolated by non-resonant dual active bridge (DAB) converter or a resonant LC, LLC, or CLLC converter [15,16]. In [17,18,19,20], a three-stage topology based on modular multilevel converter (MMC) has been proposed, which includes four ports. MMC is used to realize electrical energy conversion between the MVAC port and MVDC port, while isolated DC/DC converters with input-series–output-parallel (ISOP) topology are used to realize electrical energy conversion between the MVDC port and LVDC ports. A low-voltage inverter is used to realize electrical energy conversion between the LVDC port and LVAC port. Under the same voltage and power level, the topology based on CHB uses fewer power modules, while the topology based on MMC is more suitable for applications with MVDC port requirement.

In [21], a multi-port PET with common DC bus has been presented. The common low-voltage DC bus serves as the internal bus to realize energy exchange between the ports. This type of multi-port PET can be considered as a simple combination of converter in each port, which can be designed separately in hardware, and the control strategies between ports can also be independent of each other. However, due to the use of a large number of intermediate energy conversion links, it will increase costs and reduce efficiency.

To reduce the large number of power modules and components used in multi-port PET and avoid redundant intermediate power conversion links, a four-port PET based on multiple active bridges (MAB) converter is proposed in [22,23]. A multi-winding high-frequency transformer is adopted to connect each port, which can significantly reduce the number of electrical energy conversion links. Due to the limitations of insulation and heat dissipation processes of the multi-winding high-frequency transformers, this topology is difficult to apply in high-voltage and high-power applications. In [24], a DC PET topology is proposed, in which the ports are connected through a medium-voltage high-frequency link. The transformation from MVDC port to the medium-voltage high-frequency link is achieved through a four-bridge arm MMC. Multiple two-winding high-frequency transformers are connected in series, and the secondary sides of each high-frequency transformer are rectified by an H-bridge and connected in parallel to form an LVDC port. This type of structure can reduce the number of electrical energy conversion links. However, the locking of the faulty module affects the normal operation of other modules due to the series connection of multiple high-frequency transformers. This will limit the scalability of the structure. In [25,26], a multi-port PET topology with a common low-voltage high-frequency link has been proposed. The topology is easy to expand to the interconnection of any number of high-frequency transformers, which is easier for modular design. However, it requires high consistency in waveform amplitude and phase for each port of the common high-frequency link. The energy collection link between the ports is the high-frequency link. Due to the lack of energy storage capacitors as an energy snubber, there is strong coupling between modules and ports [26]. Therefore, whether from the main circuit hardware or the control strategy of each port, it is necessary to integrate and co-ordinate the design of multi-port PET, which undoubtedly increases the complexity of the design.

In terms of a control strategy for multi-port PET, an adaptive bidirectional droop control is proposed to maintain the DC bus voltage with less communication [27]. A communication-free edge-control strategy for energy routers based on cyber-energy dual modulation is proposed to achieve power balance without the communication network [28]. To realize power balancing, voltage balancing control based on the power decoupling calculations is adopted in [29] and an unbalanced power control strategy for a cascaded dual active bridge (DAB) converter is proposed [30]. Based on the single-phase dq model, a novel voltage and power control strategy is proposed to balance the voltages and power [31]. In [32], a hierarchical management control strategy is applied in a multi-port energy router in smart home. In [33], an isolated three-port bidirectional dc–dc converter is proposed. Duty cycle control and phase-shift control are combined to minimize the overall system losses. In order to improve the reliability of the multi-port PET, a low-voltage ride-through (LVRT) control strategy is proposed in [34].

Based on the analysis above, a topology of five-port DC PET with a common DC bus is proposed. The proposed topology in this paper is simple, efficient, and the ±375 V port can automatically achieve power imbalance output. A conversion circuit based on soft switching technology is connected in series on the 750 V DC bus of the ISOP DC transformer [35] to achieve ±375 V voltage and 300–500 V adjustable voltage port output.

Table 1. Basic characteristic comparison of different structures.

	Topological Complexity	Control Complexity	Efficiency	Scalability	Difficulty of Manufacture
Series connected type [10,11,12,13,14,15,16,17,18,19,20]	Low	Medium	Medium	Low	Low
Common DC bus type [21]	High	Low	Medium	High	Low
Common high-frequency link type [22,23,24,25,26]	High	High	Low	Medium	High
The proposed PET	Low	Low	High	High	Low

lists the comparison of different structures in terms of the topological complexity, control complexity, efficiency, scalability, and difficulty of manufacture. Firstly, the working principles of ±375 V and 300–500 V adjustable circuits are analyzed, and the parameter design method and control strategy for the circuit are proposed. Finally, the correctness and effectiveness of the proposed topology and control strategy are verified through simulation and experiments.

2. Topology of the Multi-Port DC Transformer

The proposed topology of the multi-port DC PET mainly includes five ports, as shown in Figure 1.

Port 1 (20 kV DC port): generally, 10 kV AC power is converted into 20 kV DC power through the voltage source converter.

Port 2 (750 V DC port): by adopting a voltage balance unit (VBU) and LLC resonant converter, the 20 kV DC power in port 1 is converted into 750 V DC power [27].

Port 3 (300–500 V adjustable port): the 750 V DC power in port 2 is converted into adjustable DC power ranging from 300 to 500 V through a buck converter. The buck converter can achieve high-efficiency conversion by adopting soft switching technology.

Port 4 and port 5 (±375 V DC port): the 750 V DC power in port 2 is converted into ±375 V true bipolar DC power through VBU. The VBU can achieve zero-current turn-on and turn-off by the auxiliary resonant circuit.

3. 300–500 V Adjustable Port

The 300–500 V adjustable port achieves energy exchange with the 750 V port by adopting a buck converter with soft switching. Meanwhile, the voltage and power of the port can be regulated in real time by using closed-loop control to meet the port requirements.

3.1. Working Principle of 300–500 V Adjustable Port

The interleaved buck converter (IBC) is adopted for the 300–500 V adjustable port, as shown in Figure 2. The upper bridge arm switch can achieve zero-current switch (ZCS turn-on), and the lower bridge arm diode can achieve zero-current switch (ZCS turn-off). The current equivalent frequency of the filter inductor L is twice the switching frequency.

The working waveform of the IBC is shown in Figure 3. G₁–G₄ are the driving signals of switches S₁–S₄. i_L1 and i_L2 are the currents of auxiliary inductors L₁ and L₂, and i_L is the current of filter inductor L. The reference direction of the currents is shown in Figure 2. i_s1–i_s4 are the currents of switches S₁–S₄, and the reference direction of the currents is the current direction flowing through the IGBTs (i_s1–i_s4 > 0 indicates the current flowing through the IGBTs, and i_s1–i_s4 < 0 indicates the current flowing through the anti-parallel diodes).

There are six modes in one working cycle of the IBC.

Mode 1 (t₀–t₁): before t₀, the anti-parallel diode of switch S₃ is in the off state, and the current i_L of filter inductor L flows through the auxiliary inductor L₂ and the anti-parallel diodes of switch S₄. Thus, i_L2 equals i_L. At t₀, switch S₁ is turned on, and i_L1 flows from S₁, L₁, and L to the output port. At the same time, a freewheeling circuit of i_L2 is formed through switch S₄, L₂, L, and the output port. The current commutation from i_L2 to i_L1 is completed. The current i_L2 of auxiliary inductor L₂ drops to zero at the end of the mode. Therefore, the switch S₁ can achieve ZCS turn-on, and the anti-parallel diode of switch S₄ can achieve ZCS turn-off. In this mode, S₂ and S₃ withstand the bus voltage V₇₅₀. The diagram of this stage is shown in Figure 4a.

Mode 2 (t₁–t₂): the switch S₁ remains on, and the current i_L1 of auxiliary inductor L₁ flows from the input port, S₁, L₁, and L to the output port. The current i_L1 of auxiliary inductor L₁ equals the current i_L of filter inductor L at this stage. In this mode, S₂ withstands the bus voltage V₇₅₀. S₃ and S₄ withstand the bus voltage V₇₅₀ together. The diagram of this stage is shown in Figure 4b.

Mode 3 (t₂–t₃): switch S₁ is turned off at t₂. The freewheeling circuit of i_L1 includes the anti-parallel diode of switch S₃, L₁, L, and the output port. In this mode, S₁ withstands the bus voltage V₇₅₀. S₃ and S₄ withstand the bus voltage V₇₅₀ together. The diagram of this stage is shown in Figure 4c.

Mode 4 (t₃–t₄): the switch S₂ is turned on at t₃. The current i_L2 of auxiliary inductor L₂ flows from S₂, L₂, and L to the output port. At the same time, a freewheeling circuit of i_L1 is formed through the anti-parallel diode of switch S₃, L₂, L, and the output port. The current commutation from i_L1 to i_L2 is completed. The current i_L1 of auxiliary inductor L₁ drops to zero at the end of the mode. Therefore, the switch S₂ can achieve ZCS turn-on, and the anti-parallel diode of switch S₃ can achieve ZCS turn-off. In this mode, S₁ and S₄ withstand the bus voltage V₇₅₀. The diagram of this stage is shown in Figure 4d.

Mode 5 (t₄–t₅): the switch S₂ remains on, and the current i_L2 of auxiliary inductor L₂ flows from the input port, S₂, L₂, and L to the output port. The current i_L1 of auxiliary inductor L₂ equals the current i_L of filter inductor L at this stage. In this mode, S₄ withstands the bus voltage V₇₅₀. S₁ and S₂ withstand the bus voltage V₇₅₀ together. The diagram of this stage is shown in Figure 4e.

Mode 6 (t₅–t₆): Switch S₂ is turned off at t₅. The freewheeling circuit of i_L2 includes the anti-parallel diode of switch S₄, L₂, L, and the output port. In this mode, S₃ withstands the bus voltage V₇₅₀. S₁ and S₂ withstand the bus voltage V₇₅₀ together. The diagram of this stage is shown in Figure 4f.

According to the analysis of the working principle of IBC during one cycle, it can be concluded that the upper arm of the converter can achieve ZCS turn-on and hard turn-off, and the lower arm of the converter can achieve ZCS turn-off. Therefore, the IBC has a higher efficiency due to good soft switching characteristics.

3.2. Parameter Design of 300–500 V Adjustable Port

According to the analysis of the working principle, the current i_L1 and i_L2 of the auxiliary inductors during Mode 1 satisfy Equations (1) and (2).

$${\displaystyle \frac{V_{750}-V_{\mathrm{mid}}}{L_1}}·\left(t_1-t_0\right)=I_{\mathrm{L}1},$$

(1)

$${\displaystyle \frac{V_{\mathrm{mid}}}{L_2}}·\left(t_1-t_0\right)=I_{\mathrm{L}0},$$

(2)

where I_L0 and I_L1 are the current i_L of the filter inductor L at t₀ and t₁, and V_mid is the voltage shown in Figure 4.

Since the inductance of auxiliary inductors L₁ and L₂ are the same, and I_L0 ≈ I_L1, Equation (3) can be derived as follows:

$$\begin{array}{l}{V}_{750}-V_{\mathrm{mid}}=V_{\mathrm{mid}}\\ \Rightarrow V_{\mathrm{mid}}=0.5\times V_{750}\end{array},$$

(3)

Therefore, ZCS turn-on of the switch and ZCS turn-off of the diode can be achieved by designing auxiliary inductors L₁ and L₂ to control the current rise rate of the switch and the current drop rate of the anti-parallel diode.

The current i_L of the filter inductor increases linearly in Mode 2 and can be expressed as:

$$\begin{array}{l}{I}_{\mathrm{L}2}-I_{\mathrm{L}1}={\displaystyle \frac{V_{750}-V_{300-500}}{L+L_1}}\times \left(t_2-t_1\right)\\ \Delta I_{\mathrm{Lmax}}\approx {\displaystyle \frac{V_{750}-V_{300-500}}{L+L_1}}\times D{T}_{\mathrm{s}}\end{array},$$

(4)

where D is the duty cycle of S₁ and S₂, and T_s is the switching cycle. The relationship between the input and output voltage can be expressed as:

$${\displaystyle \frac{V_{300-500}}{V_{750}}}=2D,$$

(5)

Based on the ripple of the filter inductor current, the value of the filter inductor L can be calculated by Equations (4) and (5).

Limiting the current rise rate of the switch and the current drop rate of the diode to below 20 A/μs, the auxiliary inductors L₁ and L₂ can be calculated as 20 μH through Equations (1) and (2). Limiting the current i_L fluctuation below 20%, the inductor can be calculated as 170 μH through Equations (4) and (5).

4. ±375 V Port

The ±375 V port can achieve true bipolar input or output of ±375 V voltage. The energy exchange between ±375 V port and 750 V port can be achieved through two-capacitor VBU (TC-VBU). The open-loop fixed frequency control of VBU is adopted, and the control strategy is simple.

4.1. Working Principle of ±375 V Port

The TC-VBU topology of ±375 V port is shown in Figure 5. The TC-VBU consists of two half-bridges (S_1a, S_1b and S_2a, S_2b), which are connected in series. Each half-bridge is connected in parallel with a DC-link capacitor (C₁ and C₂), and an LC resonant branch (L_p1, C_p1) is connected between the midpoints of each half-bridge to transfer energy between the DC-link capacitors. The TC-VBU can achieve voltage balance between +375 V port and −375 V port under power unbalance condition.

Before analyzing the working mode and principle, the following assumptions are made:

(1)

All components in the circuit are ideal and ignore the influence of parasitic parameters.

(2)

The switch signal of the upper and lower switch of the half-bridges are synchronized.

(3)

The resonant cycle of the resonant branch (L_p1, C_p1) is T_r. The switch cycle of the TC-VBU is T_s and the dead time of the switch is T_d. Thus, T_r = T_s − 2T_d.

The working mode and principle of the TC-VBU under unbalanced power between ±375 V ports in the forward direction are shown in Figure 6. The working waveform is shown in Figure 7.

There are four modes in one working cycle.

Mode 1 (t₀–t₁): S_1a and S_2a are turned on at t₀. The resonant capacitor C_p1 is charged by DC-link capacitor C₁ through the resonant branch, which is composed of C₁, S_1a, L_p1, C_p1, and the anti-parallel diode of switch S_2a. The resonant current i_Lp1 starts to resonate from zero in the forward direction at time t₀ and, after half of the resonant cycle, it resonates back to 0 at time t₁. S_1a and S_2a can achieve ZCS turn-on. In this mode, S_1b withstands the voltage of DC-link capacitor V₊₃₇₅. S_2b withstands the voltage of DC-link capacitor V₋₃₇₅. The diagram of this stage is shown in Figure 6a.

Mode 2 (t₁–t₂): S_1a and S_2a are turned off at t₁. At this point, the resonant current i_Lp1 is already zero. Therefore, S_1a and S_2a can achieve ZCS turn-off. In this stage, only DC capacitor C₂ provides energy to the load. In this mode, S_1a, S_1b, S_2a, and S_2b withstand the bus voltage V₇₅₀ together.

Mode 3 (t₂–t₃): S_1b and S_2b are turned on at t₁. The resonant capacitor C_p2 is charged by DC-link capacitor C₂ through the resonant branch, which is composed of C₂, S_1b, L_p1, C_p1, and the anti-parallel diode of switch S_2b. The resonant current i_Lp1 starts to resonate from zero in the backward direction at time t₀ and, after half of the resonant cycle, it resonates back to 0 at time t₁. S_1b and S_2b can achieve ZCS turn-on. In this mode, S_1a withstands the voltage of DC-link capacitor V₊₃₇₅. S_2a withstands the voltage of DC-link capacitor V₋₃₇₅. The diagram of this stage is shown in Figure 6b.

Mode 4 (t₃–t₄): S_1b and S_2b are turned off at t₃. At this point, the resonant current i_Lp1 is already zero. Therefore, S_1b and S_2b can achieve ZCS turn-off. In this stage, only the DC capacitor C₂ provides energy to the load. In this mode, S_1a, S_1b, S_2a, and S_2b withstand the bus voltage V₇₅₀ together.

4.2. Parameter Design of ±375 V Port

According to the working principle of TC-VBU, the current and voltage of the resonant branch in Mode 1 can be calculated as:

$$\left\{\begin{array}{l}{i}_{\mathrm{Lp}1}(t)=I_{\mathrm{Lp}1}\sin \left[{\omega }_{\mathrm{r}}\left(t-t_0\right)\right]\\ V_{\mathrm{Cp}1}(t)=V_1-Z_{\mathrm{r}}{I}_{\mathrm{Lp}1}\cos \left[{\omega }_{\mathrm{r}}\left(t-t_0\right)\right]\end{array}\right.,$$

(6)

where I_Lp1 is the peak current of the resonant inductor, Z_r is characteristic impedance of the resonant branch, V₁ is the voltage of capacitor C₁, and ω_r is the resonant angular frequency. They can be calculated as:

$$\left\{\begin{array}{l}{I}_{\mathrm{Lp}1}=\mathrm{\pi }{\displaystyle \frac{V_1}{2R}}·{\displaystyle \frac{T_{\mathrm{s}}}{T_{\mathrm{s}}-2T_{\mathrm{d}}}}=\mathrm{\pi }{\displaystyle \frac{I_{\mathrm{o}}}{2}}·{\displaystyle \frac{T_{\mathrm{s}}}{T_{\mathrm{r}}}}=\mathrm{\pi }{\displaystyle \frac{V_{750}}{4R}}·{\displaystyle \frac{T_{\mathrm{s}}}{T_{\mathrm{r}}}}\\ Z_{\mathrm{r}}=\sqrt{{\displaystyle \frac{L_{\mathrm{p}1}}{C_{\mathrm{p}1}}}}\\ {\omega }_{\mathrm{r}}={\displaystyle \frac{1}{\sqrt{L_{\mathrm{p}1}{C}_{\mathrm{p}1}}}}\end{array}\right.,$$

(7)

According to Equations (6) and (7), the peak current of the resonant branch (I_{Lp_max}) and the peak voltage of the resonant inductor and capacitor (V_{Lp_max} and V_{Cp_max}) can be calculated as:

$$I_{\mathrm{Lp}\_\max }=\mathrm{\pi }{\displaystyle \frac{V_1}{2R}}·{\displaystyle \frac{T_{\mathrm{s}}}{T_{\mathrm{s}}-2T_{\mathrm{d}}}}=\mathrm{\pi }{\displaystyle \frac{I_{\mathrm{o}}}{2}}·{\displaystyle \frac{T_{\mathrm{s}}}{T_{\mathrm{r}}}},$$

(8)

$$V_{\mathrm{Lp}\_\max }=I_{\mathrm{Lp}\_\max }{Z}_{\mathrm{r}},$$

(9)

$$V_{\mathrm{Cp}\_\max }={\displaystyle \frac{1}{2}}{V}_{750}+I_{\mathrm{Lp}\_\max }{Z}_{\mathrm{r}},$$

(10)

From Equations (8) to (10), it can be seen that, when the output voltage and power are determined, the current stress of the resonant inductor is determined and the voltage peak of the resonant inductor and capacitor is affected by the characteristic impedance. If the peak voltage of the resonant capacitor satisfies V_{Cp_max} ≥ V₇₅₀, it will cause TC-VBU to generate additional operating modes, and the resonant circuit generated by the additional operating modes will affect the normal operation of the converter and the implementation of ZCS. Therefore, in order to avoid additional working modes, the peak voltage of the resonant capacitor must satisfy V_{Cp_max} < V₇₅₀. By substituting this constraint into Equation (10), it can obtain:

$$Z_{\mathrm{r}}<{\displaystyle \frac{2R{T}_{\mathrm{r}}}{\mathrm{\pi }{T}_{\mathrm{s}}}},$$

(11)

According to the relationship between resonant impedance and resonant inductance Z_r = L_p1ω_r, the maximum value of the resonant inductor can be calculated as:

$$L_{\mathrm{p}1}<{\displaystyle \frac{R{T}_{\mathrm{r}}^2}{{\mathrm{\pi }}^2{T}_{\mathrm{s}}}},$$

(12)

From the above analysis, it can be concluded that a smaller resonant inductance value can keep the TC-VBU away from additional operating modes and reduce the voltage stress of the resonant inductor and capacitor. However, due to the parasitic resistance of the resonant branch, the value of the resonant inductor cannot be zero. Assuming the parasitic resistance of the resonant branch is R_p, the actual resonant frequency of the resonant branch can be calculated as:

$$f_{\mathrm{r}}={\displaystyle \frac{1}{2\mathrm{\pi }\sqrt{\frac{1}{L_{\mathrm{p}}{C}_{\mathrm{p}}}-{\left(\frac{R_{\mathrm{p}}}{2L_{\mathrm{p}}}\right)}^2}}} ,$$

(13)

To ensure that the resonant frequency of the resonant branch is not affected by parasitic resistance, the resonant parameters need to meet:

$${\displaystyle \frac{1}{L_{\mathrm{p}}{C}_{\mathrm{p}}}}>100{\left({\displaystyle \frac{R_{\mathrm{p}}}{2L_{\mathrm{p}}}}\right)}^2,$$

(14)

Therefore, the minimum value of resonant inductor can be calculated as:

$$L_{\mathrm{p}}>{\displaystyle \frac{2.5R_{\mathrm{p}}}{\mathrm{\pi }{f}_{\mathrm{r}}}},$$

(15)

From Equations (12) and (15), the value range of the resonant inductor can be obtained as:

$${\displaystyle \frac{2.5R_{\mathrm{p}}}{\mathrm{\pi }{f}_{\mathrm{r}}}}<L_{\mathrm{p}}<{\displaystyle \frac{R{T}_{\mathrm{r}}^2}{{\mathrm{\pi }}^2{T}_{\mathrm{s}}}}$$

(16)

The resonant frequency is selected as f_r = 10 kHz and the parasitic resistance of the resonant branch as R_p = 20 mΩ. Considering practical application, the output voltage is selected as V₁ = V₂ = 375 V and the power is selected as 75 kW. By substituting the parameters into (16), the range of resonant inductance values can be obtained as 1.59 μH < L_p < 18.27 μH.

5. Experimental Results

To verify the working principle, parameter design method, and control strategy of the IBC and TC-VBU, an experimental platform is constructed. The corresponding parameters of the IBC and TC-VBU are listed in

Table 2. Parameters of IBC.

Parameter	Value
Input voltage	750 V
Output voltage	300–500 V adjustable
Switching frequency	7.5 kHz
Auxiliary inductor L1/L2	20 μH
Filter inductor L	170 μH
DC-link capacitor	1 mF

and

Table 3. Parameters of TC-VBU.

Parameter	Value
Input voltage	750 V
Output voltage	±375 V
Resonant capacitor	100 μF
Resonant inductor	2.55 μH
DC-link capacitor	1 mF
Input voltage	750 V

. The power circulation method between IBC and TC-VBU is adopted. The experimental circuit is shown in Figure 8. IBC adopts a constant power control method, and TC-VBU adopts an open-loop control method.

The experimental platform is shown in Figure 9. The experimental set-up includes a programable DC power, an IBC module, an IBC control board, a TC-VBU module, a TC-VBU control board, a current/voltage sensor, control system, and monitor system. The control system and monitor system are connected via ethernet cable. The monitor system sends the working mode to the control system, while the control system uploads the module’s working status to the monitor system for display. The control system and control board are connected through optical fibers. The control system calculates the duty cycle (D_IBC) based on the difference between the reference current (I_oref) value and the measured current value (I_o) by the current sensor and sends it to the IBC control board. At the same time, the control system sends switch frequency (f_sTC-VBU) and duty cycle (D_TC-VBU) to the TC-VBU control board. The control board and the modules are connected by cables. The control board generates drive signals with dead time and drives the module switches. The closed loop control diagram of IBC is shown in Figure 10a, and the open-loop control diagram of TC-VBU is shown in Figure 10b. The models of key components in IBC and TC-VBU are listed in

Table 4. Models of key components in IBC and TC-VBU.

Key Components	Models
Switch of IBC	BSM300D12P2E001 (SiC-Mosfet from ROHM, Kyoto, Japan)
Core material of IBC inductor	Ferrite (Shijiazhuang, China)
Winding of IBC inductor	Copper wire (designed by supplier, Shijiazhuang, China)
Switch of TC-VBU	FF600R12ME4 (IGBT from Infineon, Neubiberg, Germany)
Core material of resonant inductor	Edge (from Magnetics, Pittsburgh, America )
Winding of resonant inductor	Litz wire (Yangzhou, China)

.

Before the experiment, all control boards passed an electromagnetic compatibility (EMC) test. The signal line adopts a twisted pair cable and the shorter the better.

The waveforms of IBC output voltage, output current, and filter inductor (L) current both in forward and backward mode are shown in Figure 11. The voltage of −375V port remains stable when the circulating power is 75 kW. The TC-VBU can achieve voltage balance when the output power of the two ports (+375 V port and −375 V port) is extremely unbalanced. It can be seen from the current of the filter inductor L that the frequency of the current is twice the switching frequency.

For IBC, Figure 12 shows the voltage of the switches S₁ and S₃ and the current of the auxiliary inductor. S₁ and S₃ can achieve ZCS turn-on, and the anti-parallel diodes of S₂ and S₄ can achieve ZCS turn-off. IBC has good soft switching characteristics.

For the TC-VBU, Figure 13 shows the voltage of the switches S_1a and S_1b and the resonant current. According to the working principle of TC-VBU, the zero cross-point of both the resonant current and switch current are the same. It can be seen that TC-VBU can achieve ZCS turn-on and turn-off to ensure high efficiency. During the dead time, the resonant current remains zero.

Figure 14 and Figure 15 shows the transient waveform of the experiment. In Figure 14, the reference current of IBC in forward mode increases from 100 A to 200 A. It can be seen from Figure 14a that the circulating current reaches 200 A within 10 ms. It can be seen from Figure 14b,c that the soft-switching performance during the transient state in forward mode is consistent with the steady state shown in Figure 12a and Figure 13a. In Figure 15, the input bus voltage V₇₅₀ of IBC and TC-VBU increases from 650 V to 750 V, and the reference current of IBC in backward mode remains 200 A during this process. It can be seen from Figure 15a that V₋₃₇₅ increases with V₇₅₀ and remains half of V₇₅₀. The TC-VBU has good voltage balancing characteristics. The circulating current is controlled to 200 A when V₇₅₀ reaches 750 V. It can also be seen from Figure 15b,c that the soft switching performance during the transient state in backward mode is consistent with the steady state shown in Figure 12b and Figure 13b.

Figure 16 shows the experimental efficiency curves of IBC and TC-VBU under different power points. It can be seen that both IBC and TC-VBU have high efficiency by adopting soft switching technology. The highest efficiency of IBC can reach 98.3%, and the highest efficiency of TC-VBU can reach 99.1%. The loss distribution under 75 kW is shown in

Table 5. Loss distribution under 75 kW.

Parameter	Loss Value
Mosfets of IBC	426 W
Inductors of IBC	Unknown (designed by supplier)
IGBTs of TC-VBU	513 W
Resonant inductors of IBC	45 W

. The loss of switches is calculated by PLECS simulation. The loss of resonant inductors is calculated by the parameter of core and windings. According to Table 5, the efficiency under the rated power is higher than the efficiency in Figure 16. This is due to the fact that the losses of capacitors are not included. Meanwhile, there are some errors due to the calculation method.

6. Conclusions

This paper proposes an improved multi-port DC transformer based on IBC and TC-VBU to achieve multi-voltage-level output with high efficiency. The working principles of IBC and TC-VBU are analyzed. The mathematical models in different modes are established. The control methods for the two converters are proposed. According to the mathematical models, the IBC and TC-VBU parameters are designed, including auxiliary inductor, filter inductor of IBC, resonant inductor, and resonant capacitor of TC-VBU. Simulation and experimental results verify the excellent performance of the proposed topology and the correctness of the parameter design method. Efficiency above 98% is achieved at extreme power unbalance condition. This will have a good application prospect.