Reduction in the Number of Current Sensors of a Semi-Bridgeless PFC Rectifier Based on GaNFET Characteristics

Abstract

The semi-bridgeless power factor correction (PFC) rectifier is widely used due to its high power factor, high efficiency, and low electromagnetic interference. However, in this rectifier, the inductor current will flow through the body diode of the metal–oxide–semiconductor field-effect transistor (MOSFET) when the MOSFET does not work, causing a problem in detecting the inductor current. Consequently, the current transformers are generally used as current sensors. This means that using many current sensors will make the cost and the peripheral detection circuit complicated. In this paper, our new method is to use a gallium nitride field-effect transistor (GaNFET) to replace the metal–oxide–semiconductor field-effect transistor (MOSFET) in the main switch selection. The reverse-biased conduction voltage of the third quadrant of the GaNFET is higher than the forward-biased conduction voltage of the diode, which solves the problem in detecting the inductor current, reduces the number of current sensors, and simplifies the corresponding peripheral circuits and components. Eventually, via mathematical deduction and hardware implementation, a semi-bridgeless PFC prototype with a GaNFET was built to verify the effectiveness of the proposed structure.

1. Introduction

With the rapid development of information and technology, there are more and more electronic devices for industry, business, and even the home. Cell phones, personal computers, home appliances, and other electronic products are also undergoing rapid and continuous innovation. The power supply used by these electronic devices is DC voltage, but most of the current power systems provide AC power, so it is necessary to convert the AC power to DC voltage. The simplest way to do this is to use a bridge rectifier consisting of diodes to rectify the AC power supply, which is then filtered by a bulk capacitor to obtain the DC voltage [1,2,3]. However, the input current is a steep pulse due to the charging current in the capacitor. Consequently, the current harmonics are large, causing serious harmonic pollution to the power grid and resulting in interference with the normal operation of other electrical equipment connected to the power grid. Accordingly, it is necessary to take measures to limit the current harmonics generated by these electronic devices, and the International Electrotechnical Commission (IEC) has issued the IEC 61000-3-2 current harmonic standard, which formally regulates the current harmonics created by electronic devices in detail [4,5,6].

In order to comply with the limit values of the IEC 61000-3-2 current harmonic standard, the power factor correction (PFC) must be added to make the input current sinusoidal and in phase with the input voltage of the power supply to increase the effective power so that the harmonic components of the input current will be reduced. The power factor correction (PFC) rectifier can be categorized as passive or active [7]. The traditional active PFC rectifier has a bridge rectifier at the front end, which accounts for a significant portion of the power loss. Therefore, if the power loss of this bridge rectifier can be reduced, the overall efficiency will be improved.

Figure 1 shows the basic, widely used semi-bridgeless PFC rectifier [8,9,10], which improves the overall efficiency by replacing two diodes with two MOSFETs. The disadvantages of this structure are the inconvenience of detecting the input voltage and inductor current [11,12,13] and the problem of electromagnetic interference (EMI) between the input and output due to the common-mode noise generated by high-switching power semiconductor devices [14,15].

Figure 2 shows the semi-bridgeless PFC rectifier [16,17,18,19,20,21,22,23,24,25,26,27,28,29], which is a modified version of the basic, widely used semi-bridgeless PFC rectifier. In this modified circuit, two additional slow diodes, D₃ and D₄, are connected between the negative and positive terminals of the input AC voltage v_in and the negative terminal of the output capacitor C_o, respectively, to reduce the common-mode noise and, hence, to solve the EMI problem [14,15]. Furthermore, due to these two diodes, the input voltage can be detected directly by the resistive voltage divider. However, the inductor current will flow through the body diode of the inactivated MOSFET main switch, leading to an error in detecting the inductor current, causing a more complex current-detecting circuit to be required.

In order to solve the problem of detecting the inductor current, [30] suggests three methods to obtain the detected inductor current signal CS: Hall sensor detection, differential amplifier detection, and current transformer detection. These three methods are described below:

(1) Hall effect sensor detection

This method only requires the Hall effect sensor to be placed directly on the input to detect the inductor current, as shown in Figure 3. This method is simple, and the detected value is accurate and reliable, but the Hall sensor is more expensive.

(2) Differential amplifier detection

This method is to connect a current-detecting resistor R_S in series with the input, and then to use a differential amplifier DA to detect the inductor current signal on R_S, as shown in Figure 4. This method is simple and relatively inexpensive. Since the current-sampling resistor is placed at the negative terminal of the AC input voltage v_in, the detected CS is easily interfered with by the common-mode noise, resulting in a relatively low power factor.

(3) Current transformer detection

As shown in Figure 5, this method requires the use of three current transformers; the main switch S₁ and the main switch S₂ are each connected in series with the current transformers C_T1 and C_T2, the diodes D₅ and D₆, and the resistors R₁ and R₂, respectively, whereas the output is also connected in series with the current transformer C_T3, the diode D₇, and the resistor R₃. The following is a brief description of the corresponding operating principle:

(a) When v_in > 0, the main switch S₁ is turned on, the inductor L₁ stores energy, the inductor current flows through the current transformer C_T1, and CS is detected. When v_in > 0, the main switch S₁ is cut off, the inductor L₁ releases energy through diode D₁, the inductor current flows through current transformer C_T3, and CS is detected.

(b) When v_in < 0, the main switch S₂ is turned on, the inductor L₂ stores energy, the inductor current flows through the comparator C_T2, and CS is detected. When v_in < 0, the main switch S₂ is cut off, the inductor L₂ releases energy through diode D₂, the inductor current flows through current transformer C_T3, and CS is detected.

By integrating the detected inductor current signals from the above three current transformers, these inductor current signals are then converted into voltage signals that can be used by the controller via the peripheral circuits. Therefore, this method uses a large number of current transformers and complex peripheral circuits.

In this paper, our new method is to replace the metal–oxide–semiconductor field-effect transistor (MOSFET) with a gallium nitride field-effect transistor (GaNFET) in the selection of the main switch. The reverse-biased conduction voltage of the third quadrant of the GaNFET is much higher than the forward-biased voltage of the diode, which solves the problem of detecting the inductor current, reduces the number of current sensors, and simplifies the corresponding peripheral circuits and components.

2. Operating Principle of the Semi-Bridgeless PFC Rectifier Using MOSFET Main Switches

Figure 6 shows the waveforms relevant to the circuit operation during the positive and negative half-cycles of the input AC voltage, as well as these waveforms under high-frequency switching corresponding to the peak value of the sine-wave voltage. From this figure, it can be seen that the circuit has four operating states. When the input AC voltage v_in is under the positive half-cycle, the main switch S₁ is turned on/off under high-frequency switching and the main switch S₂ is always cut off. When the input AC voltage v_in is under the negative half-cycle, the main switch S₁ is cut off and the main switch S₂ is turned on/off under a high switching frequency. There are four operating states for this semi-bridge rectifier, and they are described in the following text.

• State 1: [t₀ ≤ t ≤ t₁]

As shown in Figure 7, when the input AC voltage v_in is under the positive half-cycle, i.e., v_in > 0, the main switch S₁ conducts, the main switch S₂ is cut off, the diode D₃ conducts, and the diodes D₁, D₂, and D₄ are all cut off. The input current i_in flows through the inductor L₁, the main switch S₁, and the diode D₃, and another small portion of the current flows through the body diode of the main switch S₂ and the inductor L₂. At the same time, the voltage v_L1 across the inductor L₁ is the input voltage v_in, and the inductor L₁ is in a state of magnetization due to the positive voltage across the inductor L₁. During this state, the inductor current i_L1 rises linearly, and the inductor L₁ stores energy. The energy required for the output resistor R_o is supplied by the output capacitor C_o.

To speak more lucidly, in Figure 7, the main switch S₁ is on, so the body diode of S₂ and the diode D₃ are both forward-biased; hence, there are two inductor current paths returning to the input voltage v_in. One path has the diode D₃ connected in series with the inductor L₁, whereas the other path has the body diode of S₂ connected in series with the inductors L₁ and L₂. Accordingly, the impedance of the path with the body diode of S₂ is higher than that of the path with diode D₃, so a small portion of the inductor current will flow through the body diode of S₂, and the majority of the inductor current will flow through the diode D₃. Note that the switching frequency is much higher than the line frequency. Therefore, the voltage drop due to the line frequency is quite low, so the forward-biased voltage of the body diode of the main switch S₁ is similar to the voltage forward-biased voltage of the diode D₃, causing the small portion to seem constant.

• State 2: [t₁ ≤ t ≤ t₂]

As shown in Figure 8, when the input voltage v_in is under the positive half-cycle, i.e., v_in > 0, the main switches S₁ and S₂ are cut off, the diodes D₁ and D₃ are turned on, and the diodes D₂ and D₄ are cut off. The input current i_in flows through the inductor L₁, the diodes D₁ and D₃, the output capacitor C_o, and the output resistor R_o, and another small portion of the current flows through the body diode of the main switch S₂ and the inductor L₂. At the same time, the voltage v_L1 across the inductor L₁ is the input voltage v_in minus the output voltage V_o, and the inductor L₁ is in a state of demagnetization due to the negative voltage across the inductor L₁. During this state, the inductor current i_L1 decreases linearly and the inductor L₁ releases energy. The energy required for the output resistor R_o is supplied by the input voltage v_in and the inductor L₁, which also charge the output capacitor C_o.

• State 3: [t₃ ≤ t ≤ t₄]

As shown in Figure 9, when the input AC voltage v_in is under the negative half-cycle, i.e., v_in < 0, the main switch S₁ is cut off, the main switch S₂ conducts, the diode D₄ is turned on, and the diodes D₁, D₂, and D₃ are all cut off. The input current i_in flows through the inductor L₂, the main switch S₂, and the diode D₄, and another small portion of the current flows through the body diode of the main switch S₁ and the inductor L₁. At the same time, the voltage v_L2 across the inductor L₂ is the input voltage v_in, and the inductor L₂ is in a state of magnetization due to the positive voltage across the inductor L₂. During this state, the inductor current i_L2 rises linearly, and the inductor L₂ stores energy. The energy required for the output resistor R_o is supplied by the output capacitor C_o.

• State 4: [t₄ ≤ t ≤ t₅]

As shown in Figure 10, when the input AC voltage v_in is under the negative half-cycle, i.e., v_in < 0, the main switches S₁ and S₂ are cut off, the diodes D₂ and D₄ are turned on, and the diodes D₁ and D₃ are cut off. The input current i_in flows through the inductor L₂, the diodes D₂ and D₄, the output capacitor C_o, and the output resistor R_o, and a small portion of the current flows through the body diode of the main switch S₁ and the inductor L₁. At the same time, the voltage v_L2 across the inductor L₂ is the input voltage v_in minus the output voltage V_o, and the inductor L₂ is in a state of demagnetization due to the negative voltage across the inductor L₂. During this state, the inductor current i_L2 decreases linearly, and the inductor L₂ releases energy. The energy required for the output resistor R_o is supplied by the input voltage v_in and the inductor L₂, which charge the output capacitor C_o.

3. Current Sensor Improvement Based on a GaNFET

In the following subsections, we will mainly discuss how to reduce the number of current sensors.

3.1. Operational Characteristics of the GaNFET in the Third Quadrant

From the literature [31], it can be seen that the structure of the GaNFET has a channel that generates two-dimensional electron gas (2DEG) by connecting the source and drain electrodes, and the gate voltage is used to control the conductivity of this channel, as shown in Figure 11.

To speak lucidly, there are three operating cases for the GaNFET in Figure 11, as described below, where V_th is the threshold voltage and R_on is the channel’s turn-on resistance:

Case 1: As V_gs > V_th and V_DS > 0, the channel is turned on. The current I_DS is operated in the first and third quadrants, and this operation feature is the same as that of the MOSFET. The corresponding drain–source voltage V_DS equation, called the forward-biased conduction voltage, can be expressed as follows:

$$V_{DS}=I_{DS}\cdot R_{on}$$

(1)

Case 2: As V_gs < V_th and V_DS > 0, the channel is turned off. The current I_DS is not operated in the first quadrant, and this is the same as for the MOSFET.

Case 3: As V_gs < V_th and V_DS < 0, the channel has a chance to be turned on, with I_DS operated in the third quadrant. The corresponding source–drain voltage V_SD equation, called the reverse-biased conduction voltage, can be expressed as follows:

$$V_{SD}\approx \left(V_{th}-V_{gs}\right)+I_{SD}\cdot R_{on}$$

(2)

As compared with the MOSFET, since V_DS < 0, the body diode of the MOSFET will be turned on and operated in the third quadrant, with a forward-biased voltage of about 0.6 V to 1.5 V, but without (2). But the GaNFET has a relatively high reverse-biased conduction voltage V_SD, with the voltage V_th − V_gs typically being higher than 0.6 V when the voltage V_gs is set to zero. For example, the threshold voltage V_th of a commercially available 650 V GaNFET is about 1.7 V. Therefore, the improvement proposed in this paper is to utilize the fact that the reverse-biased conduction voltage V_SD of the GaNFET is higher than the forward-biased conduction voltage V_DS of the diode when the GaNFET is operated in the third quadrant.

The operating range of V_DS for the GaNFET is based on industrial applications. The blue line is suitable for bidirectional operation, for example, battery charging and discharging, whereas the red line is suitable for regenerative operations, for example, motor regeneration (Figure 12).

3.2. Current Flow with GaNFETs Used as Main Switches

In this paper, a method is proposed to improve the inductor current detection for semi-bridgeless PFC rectifiers by replacing the MOSFET main switches with the GaNFET main switches shown in Figure 7. As shown in [31], the reverse-biased conduction voltage V_SD of the GaNFET main switches S₁ and S₂ in the third quadrant is much higher than the forward-biased voltage of the diodes D₃ and D₄, such that there is no inductor current flowing through the inactivated main switch and the inductor. The key to this improvement lies in the fact that there is only one inductor current path at any time, and it only necessary to connect a current-detecting resistor R_S in series between the sources of the GaNFET main switches S₁ and S₂ and the anodes of the diodes D₃ and D₄, as shown in Figure 13. Accordingly, the voltage across the current-detecting resistor R_S contains information on the real inductor current. In this way, the inductor current can be accurately detected, thus reducing the number of current sensors and simplifying the corresponding peripheral circuits and components.

4. Design Considerations

In this section, the system configuration adopted, the system specifications defined, and the component specifications used will be described, along with the GaNFET main switches used.

4.1. System Configuration Adopted

Figure 14 shows the overall system configuration of the semi-bridgeless PFC rectifier. This system consists of the main power stage circuit, input voltage divider, output voltage divider, current-sensing resistor R_S, AC phase detector, gate drivers, and PFC controller.

From Figure 14, it can be seen that the output voltage is sensed by the voltage divider. Also, from this figure, it can be seen that the PFC controller adopts the average current-mode control, i.e., dual-loop control. For the voltage loop control to be considered, the output voltage signal is measured by the output voltage divider and subtracted from the voltage reference, and then the error value and the measured input voltage signal are fed into the waveform generator to create the current command value in phase with the mains voltage. For the current loop control to be considered, the inductor current signal measured by the current-sensing resistor R_S is subtracted from this command value, so as to yield the desired control force. For each switching cycle, the control system calculates the turn-on time of the main switch so that the input current will be controlled in phase with the input voltage as tightly as possible, thereby resulting in a relatively high power factor and relatively low current harmonics.

4.2. System Specifications Defined

Table 1. System specifications.

Parameter	Specification
Inductor Operation Mode	Continuous Conduction Mode (CCM)
Control Strategy	Average Current-Mode Control
Input Voltage (vin)	AC 90~264 V
Output Voltage (Vo)	DC 400 V
Rated Output Current (Io,rated)	1.5 A
Rated Output Power (Po,rated)	600 W
Switching Frequency (fs)	65 kHz
Rated Load Efficiency (η)	95%

displays the system specifications for the semi-bridgeless PFC rectifier.

4.3. Component Specifications Used

Table 2. Component specifications.

Component		Specification
GaNFETs	S1, S2	NV6128
Diodes	D1, D2	IDH10G65C6
Bridge Diodes	D3, D4	LL25XB60
Inductors	L1, L2	Inductance: 300 μH
Output Capacitor	Co	120 μF/450 V $\times$ 3
AC Phase Detector		TEA2206
Gate Drivers		UCC27424
PFC Controller		TEA2017

displays the component specifications used in the semi-bridgeless PFC rectifier.

5. Experimental Results

Some measured waveforms and data are given to verify the effectiveness of the proposed method.

5.1. Measured Steady-State Waveforms

Some steady-state waveforms were measured at rated loads under input voltages of AC 90, 115, 230, and 264 V. The input voltage and inductor current were measured as shown in Figure 15, Figure 16, Figure 17 and Figure 18. From these waveforms, it can be seen that there is no inductor current flowing through the inactivated main switch and the inductor during the time when the inductor stores and releases energy.

Next, the input voltage and input current were measured as shown in Figure 19, Figure 20, Figure 21 and Figure 22. From these waveforms, it can be seen that the input current tightly follows the input voltage, so that the power factor correction can be achieved.

After this, the gate driving signal, main switch voltage, and inductor current were measured at AC voltage peaks as shown in Figure 23, Figure 24, Figure 25 and Figure 26, and there was no voltage spike on the main switches. Therefore, from these abovementioned waveforms, it can be seen that the outputs are stable at rated loads for the whole range of input voltage.

5.2. Measured Dynamic Response Waveforms

Next, we measured some dynamic response waveforms relevant to the output current, output voltage, and input current at different input voltages due to upload and download, as shown in Figure 27, Figure 28, Figure 29 and Figure 30. The input voltages were AC 90, 115, 230 and 264 V, and the output loads were varied from 20% of the rated load to the rated load and from the rated load to 20% of the rated load. From these waveforms, it can be seen that from 20% of the rated load to the rated load, the corresponding change in output voltage is about 6.5%, with a recovery time of about 150 ms, while from the rated load to 20% of the rated load, the corresponding change in output voltage is about 4.5%, with a recovery time of about 75 ms.

From the waveforms shown in Section 5.1 and Section 5.2, it can be seen that the effectiveness of the semi-bridgeless PFC rectifier with GaNFET main switches can be verified.

5.3. Measured Data

Next, the power factor PF, total harmonic distortion THD, input power P_in, and output power P_out from the rated load to 10% of the rated load, in 10% intervals, were measured under input voltages of AC 115 and 230 V, as shown in

Table 3. Measured data under an input voltage of AC 115 V.

Output Load (%)	PF	THD (%)	Pin (W)	Pout (W)	Eff. (%)
100	0.994	6.48	624.90	602.52	96.42
90	0.994	6.92	561.05	542.10	96.62
80	0.994	7.38	497.94	481.93	96.78
70	0.993	7.92	435.19	421.69	96.90
60	0.993	8.55	372.91	361.58	96.96
50	0.993	9.28	310.96	301.55	96.97
40	0.993	10.16	249.23	241.49	96.89
30	0.992	10.80	187.21	181.24	96.81
20	0.991	12.07	125.96	121.40	96.38
10	0.974	14.80	64.63	61.31	94.86

and

Table 4. Measured data under an input voltage of AC 230 V.

Output Load (%)	PF	THD (%)	Pin (W)	Pout (W)	Eff. (%)
100	0.996	6.51	612.30	602.82	98.45
90	0.996	6.77	550.99	542.45	98.45
80	0.995	7.03	489.89	482.21	98.43
70	0.993	7.36	428.99	422.13	98.40
60	0.991	7.67	368.00	361.88	98.34
50	0.986	8.28	307.24	301.78	98.22
40	0.977	8.86	246.49	241.70	98.06
30	0.957	9.95	185.72	181.57	97.76
20	0.905	10.84	125.00	121.40	97.12
10	0.720	10.99	64.30	61.30	95.34

. In addition, based on P_in and P_out, the efficiency Eff. can be calculated. Based on Table 3 and Table 4, the curves of PF vs. load current, THD vs. load current, and Eff. vs. load current were plotted as shown in Figure 31, Figure 32 and Figure 33, respectively. According to these experimental results, under an input voltage of AC 115 V and the rated output power, the PF is 0.994 and the efficiency is 96.42%, whereas under an input voltage of AC 230 V and the rated output power, the PF is 0.996 and the efficiency is 98.45%. Accordingly, this semi-bridgeless PFC rectifier prototype possesses high PF and high efficiency.

Finally, a digital power analyzer (Chroma 66202), manufactured by Chroma Inc., Taoyuan City, Taiwan, was used to measure the harmonic values of the input currents at rated loads under input voltages of AC 115 and 230 V. The test values and the limit values of IEC61000-3-2 Class D were tabulated, as shown in

Table 5. Current harmonic limit values and test values under an input voltage of AC 115 V.

Harmonic Order	IEC 61000-3-2 Class D Limit (A)	Harmonic Test Value (A)
3	2.3000	0.1390
5	1.1400	0.1400
7	0.7700	0.1599
9	0.4000	0.1475
11	0.3300	0.1308
13	0.2100	0.1025
15	0.1500	0.0695
17	0.1324	0.0330
19	0.1184	0.0034
21	0.1071	0.0238
23	0.0978	0.0392
25	0.0900	0.0410
27	0.0833	0.0376
29	0.0776	0.0228
31	0.0726	0.0090
33	0.0682	0.0099
35	0.0643	0.0219
37	0.0608	0.0282
39	0.0577	0.0289

and

Table 6. Current harmonic limit values and test values under an input voltage of AC 230 V.

Harmonic Order	IEC 61000-3-2 Class D Limit (A)	Harmonic Test Value (A)
3	2.3000	0.1085
5	1.1400	0.0919
7	0.7700	0.0717
9	0.4000	0.0504
11	0.3300	0.0362
13	0.2100	0.0290
15	0.1500	0.0294
17	0.1324	0.0122
19	0.1184	0.0037
21	0.1071	0.0053
23	0.0978	0.0089
25	0.0900	0.0083
27	0.0833	0.0073
29	0.0776	0.0056
31	0.0726	0.0011
33	0.0682	0.0061
35	0.0643	0.0030
37	0.0608	0.0077
39	0.0577	0.0074

. Figure 34 and Figure 35 were plotted based on these two tables. From these two figures, it can be seen that the semi-bridgeless PFC rectifier prototype meets the IEC 61000-3-2 Class D harmonic standard.

5.4. Experimental Setup

Figure 36 displays a photo of the prototype for testing.

6. Conclusions

In this paper, the problem of detecting the inductor current of a semi-bridgeless PFC rectifier was improved. Firstly, we introduced the GaNFET feature that the reverse-biased conduction voltage V_SD in the third quadrant is higher than the forward-biased conduction voltage of the diode. This feature was used to improve the current-detecting circuit. Accordingly, a semi-bridgeless PFC rectifier prototype was implemented to demonstrate the effectiveness of the proposed strategy.

From the experimental results, it can be seen that under different input voltages and output power ratings, the inductor current path has only one loop and does not flow through the inactivated main switch and the inductor. It was confirmed that the proposed strategy is effective, thereby reducing the number of current sensors and simplifying the number of peripheral circuits and components.

Furthermore, under an input voltage of AC 115 V and the rated output power, the PF is 0.994 and the efficiency is 96.42%, whereas under an input voltage of AC 230 V and the rated output power, the PF is 0.996 and the efficiency is 98.45%. This confirms that the semi-bridgeless PFC rectifier prototype with GaNFET main switches possesses high PF and high efficiency. Moreover, the IEC61000-3-2 Class D current harmonic standard can be met under both of these two cases.